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Low Cost Electronic Ballast for a 36-W Fluorescent Lamp Based on a Current-Mode-Controlled Boost Inverter for a 120-V DC Bus Power Distribution Manuel Rico-Secades, Member, IEEE, Emilio L. Corominas, Member, IEEE, Jorge García, Student Member, IEEE, Javier Ribas, Member, IEEE, Antonio J. Calleja, Member, IEEE, J. Marcos Alonso, Senior Member, IEEE, and Jesús Cardesín, Member, IEEE

Abstract—A non-resonant electronic ballast based on one power switch and on one reactive element-one inductance-is described in this paper. The special current mode control implementation provides an intrinsic short-circuit protection and a very simple control circuitry. Filament heating time, current during the heating period, and protection against broken tube can be easily implemented with the proposed circuitry. A greenhouse application with a 120-V dc bus power distribution is presented in this paper. The dc bus voltage is easily obtained (in this particular application) from a classical series-connected 12-pulse rectifier in combination with a transformer with two secondary windings (one in Wye connection and other one in Delta connection). An additional advantage of this configuration is the high power factor obtained at the input stage.

Fig. 1. Global view of the greenhouse lighting structure.

Index Terms—Boost inverter, current mode control, electronic ballast, fluorescent lamp, greenhouse lighting.

I. INTRODUCTION

I

N SPECIFIC applications such as greenhouse lighting with large number of lighting points, critical ambient conditions (i.e., humidity) and accessibility problems, some design considerations must be taken into account: simplicity, low cost, and safety operation. The proposed lighting system has been designed for a huge number of lamps. The prototype built (greenhouse application) is composed by 7500 lighting points distributed along a large warehouse. Critical ambient conditions (mainly humidity) imply low voltage distribution. Therefore, traditional ac distribution or typical “multilamp” strategies are not applicable [8]–[11]. In these references, are focus to multilamp systems with lamp and inverter physically closer. In the literature, ballast for large number of lamps can be found, for instance [9] (10–30 lamps nearly of the main inverter). This work proposed a different solution based on a nonresonant ballast with current control in order to achieve arc stabilization in a different way from above mentioned reference. A global view of the greenhouse lighting structure is shown in Fig. 1. The use of a common three-phase transformer and rectifier, in order to generate the dc bus voltage for the whole lighting installation, is a very useful solution.

Manuscript received April 28, 2004; revised November 12, 2004. Recommended by Associate Editor R.-L. Lin. The authors are with the UNIOVI-GEI Group, Universidad De Oviedo, Gijón E-33204, Spain (e-mail: [email protected]). Digital Object Identifier 10.1109/TPEL.2006.876821

Fig. 2. Series-connected 12-pulse rectifier.

With this solution the load is always balanced. The use of rectifier and power factor correction stage can be avoided for each electronic ballast circuit. Therefore, the cost of the whole installation can be reduced. In addition, the ripple of the bus voltage is very low (due to the 12-pulse rectifier). A classical three-phase transformer (with two secondary windings in Wye–Delta connection) and a series connection of the two rectifiers is shown in Fig. 2. The middle point of the series connection of the rectifiers is connected to ground in order to improve the installation safety. Actually, the 120-V dc bus is a 60 V/ 60 V distribution. Theoretical waveforms of phase voltage and current are shown in Fig. 3. The circuit shows a high power 0.989 and a low total harmonic distortion factor 14.17 . A small filter is enough to verify the international normative [1], [2]. The paper is focused in describing a simple electronic ballast for a fluorescent lamp for this particular application. The proposed circuit is a very simple non resonant solution. The proposal is derived from theoretical studies and analysis developed in [4]. Details about the behavior of a fluorescent lamp in steady

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(6) (7) (8) (9) The time constant of the circuit can be obtained from the expression

(10) Fig. 3. Theoretical waveforms in a phase of the power line.

Thus, an alternate voltage waveform is obtained across the lamp (the arc voltage of a 36-W fluorescent lamp is around 120 V). As the lamp is placed in parallel with the inductor the inverter cannot apply a dc voltage level to the lamp in steady-state operation. The Lamp Crest Factor used in the design is 1.4, in between the ideal value of a full square wave (1.0) and the sinusoidal waveform (1.42). Lamp crest factor is, in any case, below the recommended value 1.7. B. Ignition Basics Fig. 4. Power structure of the Boost Inverter.

state, during lamp ignition and warm-up recommendations can be found in [6] and [7]. Also, an application of this work to supply HID lamps can be found in [5]. II. THEORETICAL STUDY A. Steady State The power structure of the proposed inverter is shown in Fig. 4. The circuit operation will be described in the next paragraphs. When the power switch is ON, the 120-V bus voltage is applied to the fluorescent lamp. Simultaneously, the current through inductor (L) increases linearly

During ignition process, the fluorescent lamp behaves like an open circuit. Therefore, during the OFF stage, the inductor current circulates through the intrinsic Zener diode of the power metal–oxide–semiconductor field-transistor (MOSFET) (800 V in the design example). The ON stage is similar to the steady state operation. During this period the current in the inductance (L) increases linearly, but the lamp current is close to zero. Then both switch current and inductance current are identical During ON state '' `` (11) ``No lamp current''

(12) (13) (14)

During ON state ``Steady State Operation''

(15) (1)

During OFF state '' ``

(2)

(16)

(3)

(17)

(4)

``Through Zener diode''

(18)

``High voltage in the lamp''

(5)

(19)

When the power switch is OFF, the current through inductor flows through the fluorescent lamp (steady state operation)

A high voltage is applied to the lamp simultaneously with a controlled heating current through the filament tube. Thus, the circuit is self-protected against no lamp connection, because if the fluorescent lamp is not present, there is no current in the power elements.

During OFF state ``Normal Steady State Operation''

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applied to the filament during both steady state operation and ignition process. D. Design Process

Fig. 5. Basic control circuit based on open collector comparators.

The complete design process can be found in [3] and [4]. A detailed design process is complex. In [4], a Mathematica design program has been developed. But the complete description of this design program is out of the scope of this paper. A simplified design can be done assuming current ripple null in the inductance and duty cycle equal 0.5 [5]. Equations of this basic design have been summarized in the next lines. For an initial approach the steady state current across the inductance is assumed constant (20) Equation (20) means the current across the inductor is equal to current across the lamp. Then, the power in the lamp can be estimated in a simple way

(21) In the same way, for a specified switching frequency value can be obtained from

the

(22) Fig. 6. Basic waveforms in steady state operation.

The ripple in the inductance can be easily obtained from design equations

C. Control Circuit A very simple and cheap current mode control has been implemented in order to drive the power structure. The basic control circuit is shown in Fig. 5. Basic waveforms in steady state operation have been included in Fig. 6. The power MOSFET in maintained in ON state until the curvalue). rent through it reaches a specified value ( The current through the power MOSFET is sensed using a , and the maximum current switch simple resistor ( ) is established using an external reference ( control). A simple open collector comparator is used for this purpose (IC2). The OFF stage in the power MOSFET starts at this point. is discharged using the Simultaneously a capacitor above mentioned comparator. Once the capacitor is discharged and the switch current falls to zero, the state of IC1 changes is charged again. again, and The OFF time ( value) of the power MOSFET can be easily adjusted with the modification of the limit in the voltage capacitor ( control). across the Another external reference, and a second open collector comparator, are used for this purpose (IC1). and control is easily implemented in this So, a way. Note that an intrinsic short-circuit protection is present in the circuit. In addition, a controlled and limited heating current is

(23)

(24) Design has been done assuming the next initial values

These obtained values, are initial valid approaches to the correct ones Inductance Maximum current in the switch Switch off time Final values have been adjusted using a Mathematica program [4]. In this particular design, the following values for main elements of both circuit and control parameters have been obtained and used: Inductance Maximum current in the switch Switch off time

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Fig. 7. Stability of the operating point with fluorescent lamp variations. Fig. 9. Duty cycle versus lamp resistance.

Fig. 10. Stability of the operating point with inductance variations during manufacturing process. Fig. 8. Switching frequency versus lamp resistance.

Notice how these values match satisfactorily with the initial apvalue had to be slightly deproaches related before. The creased because the simplified design provides a higher output power to the lamp. Figs. 7–9 show the actual power-versus-load inverter behaviour and, simultaneously, the fluorescent lamp characteristic over the same diagram. These diagrams have been obtained using the mathematical program mentioned above

(25) Note that, both lamp voltage and lamp current are function of the control parameters ( and ), the and the lamp resiscircuit elements (L), the bus voltage tance . In a similar way, the fluorescent lamp characteristic has been modeled as follows [3]:

(26)

For a 36-W fluorescent lamp the constant values in the model are

The operating point can be obtained as the intersection of both characteristics. The stability of the operation point with fluorescent lamp variations is shown in Fig. 7. In the theoretical design, the operating point has been established in order to deliver 36 W in the lamp. The inverter with this control mode behaves like a power source. Therefore, fluctuations in lamp behaviour (from new lamp to old lamp) imply low fluctuation in the power level (only a few Watts) (see Fig. 7). Switching frequency and duty cycle stability again lamp resistance have been obtained and represented in Figs. 8 and 9. The stability of the operating point with inductance variations, due to manufacturing process (variation of 10%), and with the dc bus voltage fluctuation (variation of 10 %), are shown in Figs. 10 and 11. With the considered fluctuations, the power delivered to the lamp is modified in 1–2 W around the operating point.

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only during the OFF stage. As it has been stated previously, in the ignition process, the ON stage is similar to the steady state operation. Also the current in the inductance (L) increases linearly, but the lamp current is close to zero. The designed current during warm-up stage is low (only 190 mA) and the ignition voltage depends on the parasitic MOSFET Zener diode and bus voltage

Fig. 11. Stability of the operating point with dc bus voltage fluctuation.

Fig. 12. Theoretical waveforms during ignition process (not in scale).

E. Ignition Process Detail and Comments The waveforms during ignition phase are shown in Fig. 12. This figure is not scaled, and it is only intended to explain quantitatively the role of the parasitic MOSFET Zener diode during warm-up. Duty cycle and switching frequency during warm-up are different to the steady-state ones. Experimental ignition waveforms have been included in Fig. 16. Two different comments can be done about the current imposed for the ballast across the fluorescent lamp filament. During warm-up stage, the lamp filament is only heated during the MOSFET ON stage; the ignition voltage is applied

Filament current can be increased easily modifying the reference in the control circuit during the warm-up stage. Also the ignition voltage can be modified with the parasitic MOSFET Zener voltage value. The OFF stage is different. The lamp behaves like an open circuit. Thus the filament heating current flows trough the intrinsic Zener diode of the MOSFET. The prototype has been tested (several samples) using ON–OFF cycles of 3 s ON and other 3 s OFF. The filament in the tested lamp broke after 3000–4000 ON–OFF cycles. The circuit performs alike a resonant circuit without filament heating (Instantaneous ignition). A drawback of this circuit happens during ON period of steady state operation, due to the additional filament current across the filament derived from the placement of the inductor. This situation is similar in a resonant ballast (with the additional current across the parallel capacitor) but in this case, this situation takes place only during half a cycle of operation. In any case, no significant life reduction has been observed over the installation for this reason. An improvement of this ballast can be done using an external filament heating circuit, and also avoiding the circulation of the inductor current across the lamp filament in steady-state. This improvement implies additional circuit complexity and it needs the inclusion of an additional no-lamp protection circuit. The filament current is one of the drawbacks in the behaviour of this simple and economical circuit. The intrinsic Zener diode of the power MOSFET plays an important role during filament heating time. The filament heating current is maintained constant due to the implemented control method. The heating time finishes when the ignition voltage established with the Zener diode is reached, due to heating current maintained in the lamp. Different heating times can be obtained by changing the Zener voltage or/and the heating current value. With Zener voltage values below 600 V, the heating time is maintained indefinitely, and the ignition period never finishes. A no ignition protection circuitry has been included in order to limit this time to 1.5 s. Without this protection circuitry, the destruction of the power MOSFET happens after 1–2 min without ignition. The power MOSFET transistor selected (Thomson BUK 456–800-A), has been satisfactorily tested in the laboratory, in order to support the additional power during warm-up. Power MOSFET of 600, 800, and 1200 V have been tested during laboratory work with different results. In one of the tests done, using the 600-V version of this MOSFET, the warm-up period can be maintained until actuation of no-lamp protection circuit, without avalanche problems in the MOSFET transistor. The

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Fig. 13. Detail of the no ignition protection circuitry. Fig. 15. Experimental lamp voltage, current and power obtained on the prototype.

Fig. 14. Single switch inverter ballast prototype.

final choice was a power MOSFET with an intrinsic Zener diode of 800 V. A conventional glow to arc evolution follows this heating period, until the steady state operation is reached (see Fig. 10).

Fig. 16. Lamp voltage during ignition process.

F. Protection The circuit has intrinsic protections against short circuit and against tube absence. Also, if a lamp filament is broken, the oscillation of the inverter stops in a natural way. There is only one case in which the circuit is not self-protected: a broken lamp (or faulty lamp) with intact filament operation. In this case, the heating period is maintained continuously with an important power quantity dissipated in the intrinsic Zener diode (with a sure switch destruction). A very simple no ignition circuit protection has been incorporated in the ballast and it is shown in Fig. 13. If the Zener voltage is maintained during 1.5 s, a small SCR capacitor and therefore, the OFF time short-circuits the lasts indefinitely (the circuit oscillation stops until the system re-connection). A simple peak detector and a DIAC have been used for this purpose. III. PROTOTYPE A complete greenhouse system has been built and tested, following the design rules above mentioned. The photography of the proposed inverter ballast is shown in Fig. 14.

A BUK 456–800-A Power MOSFET from Thomson has been selected as a power switch. A classical E20 core with N27 material from SIEMENS has been used to implement the inductance 8.9 mH). of the circuit ( The control circuitry has been implemented using conventional components and comparators (LM393). The main characteristics of the circuit are as follows. Input Voltage: Input voltage Ripple of the input voltage

%

Lamp Parameters (Measured Over the Prototype): Fluorescent lamp Philips TLD Heating current RMS Lamp voltage Maximum Lamp voltage Lamp Crest factor RMS Lamp current Output power

RICO-SECADES et al.: LOW COST ELECTRONIC BALLAST FOR A 36-W FLUORESCENT LAMP

Fig. 17. Possible circuit improvement using and external filament heating circuit.

Circuit Performance: Input Power Efficiency % Switching frequency Duty Cycle The concordance between theoretical design and experimental measurements is really very good. This concordance is due to friendly input voltage values and circuit simplicity. The experimental lamp voltage, current and power in steady state operation are shown in Fig. 15. Again, it is important to emphasize that the voltage applied to the lamp has no dc level. As the lamp is in parallel with the inductor, the inverter cannot apply a dc voltage level to the lamp in steady-state. Finally, the experimental lamp voltage during ignition is shown in Fig. 16. During ignition process, all the current stored in the inductor (0.84 A) is dissipated in the 800 V parasitic zener diode of the MOSFET. The current discharges linearly during 9.3 S [estimated from (11) to (19)]. During the rest of the time (design value of 19.8 S), the inductor current is zero. During time can be easily estimated (from above menignition, the tioned equations) assuming a zero initial value and a linear inestimated time is around crease of the inductor current. The 62 S. A theoretical estimated period of 81.8 S is coherent with measurements obtained in Fig. 16. With this values, an estimated average power of 37 W needs to be dissipated during ignition in parasitic MOSFET zener diode. Normal ignition happens more or less in 800 S. During this time, the parasitic MOSFET zener diode must support this 37-W peak of power. The no-lamp protection circuit has been calibrated to disconnect the control if lamp does not ignite in 1.5 s. In laboratory experiments with broken lamps, this condition has been satisfactorily tested. Without any lamp protection circuitry, the destruction of the power MOSFET takes place after several minutes without ignition. IV. CONCLUSION A non-resonant electronic ballast based on one power switch and one reactive element—one inductance—has been described in this paper.

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The special current mode control implementation provides an intrinsic short circuit protection and a very simple control circuitry. Filament heating time, current during heating period and the protection against broken tube can be easily implemented with the proposed circuitry. In addition, high stability of the delivered power to the lamp, with dc bus voltage fluctuations, dispersion in fluorescent lamp characteristics and inductance variations, has been theoretically obtained and experimentally verified. A typical greenhouse application with a 120-V dc bus power distribution has been shown in this paper. A classical series-connected 12-pulse rectifier, in combination with a transformer with two secondary windings (one in Wye connection and other one in Delta connection), have been used in this application. By means of this configuration, high power factor and an adequate dc bus voltage value can be easily obtained. The main limitation is the excessive filament current during ON switching period. But in any case, no significant life reduction has been observed on the tested prototypes. A possible improvement of this circuit is suggested for future works in Fig. 17. REFERENCES



[1] Limits for harmonic current emissions (equipment input current 16 A per phase), IEC 61000-3-2 Electromagnetic Compatibility (EMC), 2005. [2] Limits for harmonic current emissions (equipment input current 16 A), IEC 61000-3-4 Electromagnetic Compatibility (EMC), 2005. [3] E. L. Corominas, J. M. Alonso, A. J. Calleja, J. Ribas, and M. Rico Secades, “Analysis of tapped-inductor inverter as low-power fluorescent lamp ballasts supplied from a very low input voltage,” in Proc. 30th Annu. IEEE Power Electron. Spec. Conf. (PESC’99), 1999, pp. 1103–1108. [4] E.L. Corominas, “Alimentación de Lámparas Fluorescentes Compactas Desde Muy Baja Tensión: Aportaciones a la Optimización del Sistema Electrónico,” Ph.D. dissertation, Univ. Oviedo, Gijon, Spain, 1999. [5] J. G. Garía, “Balastos Electrónicos No Resonantes Para Lámparas de Alta Intensidad de Descarga: Aportaciones en el Circuito de Arranque y en Las Etapas de Calentamiento y Régimen Permanente,” Ph.D. dissertation, Univ. Oviedo, Gijon, Spain, Jul. 2003. [6] C. Meyer and H. Nienhuis, Discharge Lamps. Eindhoven, The Netherlands: Philips Technical Library, 1988. [7] J. R. Coaton and A. M. Marsden, Lamps and Lighting. New York: Wiley, 1997. [8] M. A. D. Costa, M. L. Landerdahl Jr., and R. N. do Prado, “Independent multi-lamp electronic ballast,” in Proc. 37th IAS Annu. Meeting, Oct. 2002, vol. 2, no. 13–18, pp. 1065–1070. [9] R. Gules, I. Barbi, and E.M. Simoes, “A 1.2 kW electronic ballast for multiple lamps, with dimming capability and high-power-factor,” in Proc. Appl. Power Electron. Conf. Expo. (APEC’99), Mar. 1999, vol. 2, pp. 720–726, 2. [10] F. T. Wakabayashi and C. A. Canesin, “Dimmable electronic ballast with high power factor SEPIC preregulator, for multiple tubular fluorescent lamps,” in Proc. Power Electron. Spec. Conf. (PESC’04), Jun. 2004, vol. 5, no. 20–25, pp. 4043–4049. [11] T.-F. Wu, Y.-C. Liu, and Y.-J. Wu, “High-efficiency low-stress electronic dimming ballast for multiple fluorescent lamps,” IEEE Trans. Power Electron., vol. 14, no. 1, pp. 160–167, Jan. 1999.



Manuel Rico-Secades (M’88) was born in Oviedo, Spain, in 1961. He received the M.Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Gijón, Spain, in 1986 and 1989, respectively. Since 1986, he has been with the Electrical and Electronic Department, University of Oviedo, where he is currently a Full Professor. His research interests include industrial electronics and power electronics, especially resonant converters, electronics ballast, discharge lamp modeling, dc-to-dc converters, power factor correction topologies and industrial control.

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Dr. Rico-Secades received the IEEE Industrial Electronics Society Meritorious Paper Award for 1996. He is currently a member of the UNIOVI-GEI Group (Evaluated as Group of excellence by “Principado de Asturias” in Spain).

Emilio L. Corominas (M’97) was born in Oviedo, Spain, in 1965. He received the M.Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Gijón, Spain, in 1992 and 1999, respectively. In 1993, he joined the Electrical and Electronic Department, University of Oviedo, where he is currently an Associate Professor. His research interests include high-frequency electronic ballasts, discharge lamp modeling, high-frequency switching converters, power factor correction converters, and industrial control systems.

Jorge García (S’01) was born in Madrid, Spain, in 1975. He received the M. Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Gijón, Spain, in 2000 and 2003, respectively. In December 1999, he became a Researcher in the Electrical and Electronic Engineering Department, University of Oviedo, developing electronic systems for lighting and electronic switching power supplies. Since 2002 he has been an Assistant Professor of electronics. His research interests include dc/dc converters and PFC stages, switching power supplies, HF inverters for discharge lamps, and electronic starters for HID lamps.

Javier Ribas (S’97–M’04) was born in Milwaukee, WI, in 1971. He received the M.Sc. and Ph.D. degrees from the University of Oviedo, Gijón, Spain, in 1995 and 2001, respectively. In 1996, he became an Assistant Professor with the University of Oviedo and since 2002 has been an Associate Professor. His research interests include dc/dc converters, electronic lighting systems, switching power supplies, inverters, and high-power-factor rectifiers.

Antonio J. Calleja (S’96–A’98–M’04) was born in León, Spain, in 1964. He received the B.Sc., M.Sc, and Ph.D. degrees from the University of Oviedo, Gijón, Spain, in 1987, 1995, and 2000, respectively. Since 1995, he has been an Assistant Professor at the University of Oviedo. His research interests are switching-mode power supplies, high-power-factor rectifiers, high frequency electronic ballast, and ozone generation systems. Dr. Calleja is a member of the International Ozone Association (IOA).

J. Marcos Alonso (S’94–A’95–M’98–SM’03) received the M.Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Gijón, Spain, in 1990 and 1994, respectively. From 1990 to 1999, he was an Assistant Professor with the Electrical and Electronic Department, University of Oviedo, where since 1999 he has been an Associate Professor. He is the primary author for more than 40 journal and international conference papers in power and industrial electronics, and has co-authored more than one hundred. He holds four Spanish patents with one under review. His research interests include high-frequency electronic ballasts, discharge lamp modeling, power factor correction topologies, high frequency switching converters, and power converters for electrostatic applications and industrial control systems. Dr. Alonso received the IEEE Industrial Electronics Society Meritorious Paper Award in 1996. He is an active member of the Institute of Electrical and Electronics Engineers (IEEE), where he usually collaborates as transactions paper reviewer, conference session chairman, among other positions. Since October 2002 he has served as an Associate Editor of the IEEE TRANSACTIONS ON POWER ELECTRONICS in the field of Lighting Applications. He is presently serving as a Guest Editor for the Special Issue on Lighting Applications, to be published in the IEEE TRANSACTIONS ON POWER ELECTRONICS in May 2007. He is also a member of the International Ozone Association (IOA).

Jesús Cardesín (S’01–A’03–M’04) was born in Oviedo, Spain, in 1970. He received the M.Sc. and Ph.D. degrees from the University of Oviedo, Gijón, Spain, in 1995 and 2002, respectively. In 1999, he joined the Electrical and Electronic Department, University of Oviedo, where he is currently Assistant Professor. His research interests include dc/dc converters, electronic lighting systems, switching power supplies, inverters, and high-power-factor rectifiers.