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A 4.3-GHz VCO with 2-GHz Tuning Range and Low Phase Noise Paavo Väänänen, Mikko Metsänvirta, and Nikolay T. Tchamov

Abstract—A voltage-controlled oscillator (VCO) based on double cross-coupled multivibrator structure with a center frequency of 4.3 GHz and a tuning range of 2 GHz has been designed and implemented in standard 0.35- m BiCMOS technology. The measured phase noise is 113 dBc/Hz at 600-kHz offset frequency from the carrier. The VCO draws 14.6 mA from the 2.5-V supply. Index Terms—BiCMOS technology, multi-GHz VCO, wide tuning range.

I. INTRODUCTION

T

HE voltage-controlled oscillator (VCO) is one of the most challenging parts of the radio transceiver to integrate using relatively cheap BiCMOS or CMOS processes. One reason for values of the integrated inductors are low this is that the compared to those of the discrete ones. This causes, for instance, lowering of the phase-noise performance of the resonance-tank-based oscillators. Due to the switching-based principle of the relaxation oscillators the of the inductors does not affect the performance, and low quality factor integrated inductors can be used to increase the switching speed of the transistors in the circuit. This paper introduces a new version of a double cross-coupled VCO [1], [2]. Also, in this newer topology, the need for differential control is avoided, and so the control block is simplified and the tuning sensitivity can also be more easily adjusted. The VCO controlling principle is presented first. Second, the effect of using inductive loads is analyzed, the simulation and measurement results are given, and brief comments are made. Finally, the measured results are compared to other published relaxation VCO results. Implementation was carried out using the ST-Microelectronics 0.35- m BiCMOS process.

Fig. 1.

Schematic of the new double cross-coupled multivibrator ICO.

Transistors and direct the current drawn by the conwhich defines the current that bytrollable current sink passes the timing capacitor . This method requires only one controllable current sink and avoids the need for two differentially controlled sinks, as in [1], [2]. III. USING

II. VCO CONTROL PRINCIPLE The schematic of the new double cross-coupled current-controlled oscillator (ICO) is presented in Fig. 1. The current which is through the circuit is defined by the current and form the emitter-coupled kept constant. Transistors and act as a differential pair directing the multivibrator, control current through the timing capacitor [1], [2], and and together with the MOS transistors and function as active-pull-down output buffers [3].

THE

RL-LOAD IN DOUBLE CROSS-COUPLED MULTIVIBRATOR

Bandwidth extension of the amplifier by adding a zero is [4] known also as shunt peaking. Here it is used to extend the bandwidth of the multivibrator collector load circuit. is the load resistor, is the In the following equations, and equivalent capacitance as seen at the collector nodes of , and is the inductor value. Calculating the impedance of the collector load circuit when only the resistor is used gives (1)

Manuscript received June 13, 2000; revised September 8, 2000. This work was supported by the National Technology Agency of Finland (Tekes) as part of the project Digital and Analog Techniques in Flexible Receivers. P. Väänänen and M. Metsänvirta are with Nokia Mobile Phones, 33720 Tampere, Finland (e-mail: [email protected]; mikko.metsanvirta@ nokia.com). N. T. Tchamov is with the Telecommunications Laboratory, Tampere University of Technology, 33720 Tampere, Finland (e-mail: [email protected]). Publisher Item Identifier S 0018-9200(01)00444-9.

and (2) when the additional inductor is used.

0018–9200/01$10.00 © 2001 IEEE

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TABLE I SUM UP OF THE EXTRACTED SIMULATION RESULTS AND MEASUREMENT RESULTS

Fig. 3.

Differential pulse fronts.

Fig. 2. Load impedances as the function of the frequency.

Using the values , nH, and pF, and plotting the impedances as a function of the frequency produces the curves shown in Fig. 2. We can clearly see the bandwidth extension caused by adding an inductor into the circuit, and so one zero into the equivalent collector impedance function. In the time domain, this means that the rise time of the pulse is decreased. This can be analyzed easily by finding the step response of the load circuitry. The equations to be solved are simple first- and second-order differential equations and the step responses using the values mentioned above are plotted in Fig. 3. It is clear that the rise time is reduced when the inductor is used and so the circuit can be tuned into higher frequencies than when using a simple resistive load. The phase noise in relaxation oscillators is mainly caused by the uncertainty in the triggering moment [5]. The voltage over the timing capacitor that sets the relaxation period is modulated by the noise contributed by the control block and the oscillator components. Therefore, the low of the inductor is not an issue in this instance. IV. SIMULATION AND MEASUREMENT RESULTS The simplified schematic of the designed and implemented circuit is shown in Fig. 4. The circuit was simulated using CADENCE Spectre RF circuit simulator. The simulated center frequency is 4.3 GHz with a tuning range of 1.6 GHz. The simulated current consumption was 15.5 mA from the 2.5-V supply. The simulated tuning curve together with measured

one is shown in Fig. 5. The measured spectrum at 5.17 GHz with second and third harmonics is shown in Fig. 6. All the measurements were carried out using a 2.5-V supply and the measured direct current drawn from the power supply was 14.6 mA. The phase noise at the control voltage of 2.5 V is shown in Fig. 7. In Table I, the results from the extracted simulations are compiled to summarize the achievements. The chip micrograph is shown in Fig. 8. The size of the chip mm . including the pads is The measured center frequency is 4.4 GHz and the tuning range is 2 GHz. As can be seen from Fig. 5, the measured VCO tuning range is slightly greater than the simulated one since the circuit operated at lower control voltages. It can also be seen from Fig. 5 that the circuit tuning is somehow stepwise and not continuous as shown by the simulations. One possible reason for this is that the buffer in the circuit is in fact part of the oscillator core and so the core is not separated from the load circuitry. This means that all kinds of noise sources, for example transmission line reflections coming from the connections to the measurement units, can be seen at the base of the lower cross transistors, thus modulating the voltage over the timing capacitor. Additional buffers would help alleviate this problem. Another possible reason for the stepwise tuning is that either the control circuit or the control principle does not function well as in simulations. The phase-noise performance was measured using a downconverter and phase-noise measurement system. The simulator prediction was accurate close to the center frequency, e.g., at 1.4-V control voltage. In the ends of the curve, the performance

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Fig. 4. Simplified schematic of the implemented and designed double cross-coupled multivibrator VCO.

Fig. 5. Tuning curves.

was opposite to what was simulated. The difference in the low end can be explained by problems with the control of the VCO mentioned above. However, the measured output power at fundamental frequency was behaving as in the simulations, i.e., the power decreased as a function of frequency. Therefore, there is no simple explanation for failure of the simulator to predict the phase-noise performance of this circuit, but it sets a need for further investigations.

Fig. 6. The measured spectrum at 5.17 GHz. The second and third harmonics are 18.4 and 20.7 dB below the carrier frequency, respectively.

V. COMPARISON Table II presents some published relaxation oscillators [6]–[8] and two LC-resonator-based VCOs [9], [10]. In the fourth column of the table, the reported phase noise is given and in the next column, the reported phase noise is normalized

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TABLE II COMPARISON TABLE

[11]. In the class of relaxation oscillators, the phase-noise performance reported here is the best with low power consumption. Compared with the LC-tank-based VCOs, which have center frequencies comparable to our design and a broad tuning range, the VCO designed in this work has the lowest phase noise and largest tuning range with only slightly higher current consumption. VI. CONCLUSION

Fig. 7. I

Measured phase noise at a control voltage of 2.5 V, V

= 14:6 mA.

= 2 5 V, and :

A 4.3-GHz double cross-coupled VCO with 15-mA power consumption from a 2.5 -V power supply and 2-GHz tuning range was implemented in ST Microelectronic’s 0.35- m BiCMOS process. The measured phase noise is 113 dBc/Hz at 600-kHz offset from the carrier, which shows that the double cross-coupled architecture can achieve low phase noise and a large tuning range when compared to other published VCOs. A new way to control a double cross-coupled oscillator was described and the effect of using an inductive load in the collectors was analyzed to show its advantages. REFERENCES

Fig. 8. Chip micrograph. The size is 1:25

2 121 0 mm . :

to the frequency offset of 600 kHz. We have assumed that the phase noise decreases with a constant slope of 20 dB/decade

[1] N. Tchamov, A. Popov, and P. Jarske, “VCO with double cross-coupled high-speed and low-power multivibrator architecture,” in Proc. ISCAS, vol. 1, 1996, pp. 45–48. [2] N. Tchamov and P. Jarske, “Emitter-coupled multivibrator circuit,” U.S. Patent 5 825 256, Oct. 20, 1998. [3] K. M. Sharaf and M. I. Elmasry, “Active-pull-down nonthreshold logic BiCMOS circuits for high-speed low-power applications,” IEEE J. Solid-State Circuits, vol. 30, pp. 691–695, June 1995. [4] T. H. Lee, The Design of CMOS Radio-Frequency Integrated Circuits. Cambridge, U.K.: Cambridge Univ., 1998, ch. 8. [5] M. H. Wakayama and A. A. Abidi, “A 30-MHz low-jitter high-linearity CMOS voltage-controlled oscillator,” IEEE J. Solid-State Circuits, vol. SC-22, pp. 1074–1081, Dec. 1987. [6] T. Sowlati and M. H. Shakiba, “Automatic swing control in relaxation oscillators,” IEEE J. Solid-State Circuits, vol. 33, pp. 1979–1986, Dec. 1998. [7] M. Soyuer and J. D. Warnock, “Multigigahertz voltage-controlled oscillators in advanced silicon bipolar technology,” IEEE J. Solid-State Circuits, vol. 27, pp. 668–670, Apr. 1992.

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[8] B. Razavi, “A study of phase noise in CMOS oscillators,” IEEE J. SolidState Circuits, vol. 31, pp. 331–343, Mar. 1996. [9] T.-P. Liu, “A 6.5-GHz monolithic CMOS voltage-controlled oscillator,” in Proc. ISSCC, 1999, pp. 404–405.

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[10] C. Lam and B. Razavi, “A 2.6-GHz/5.2-GHz CMOS voltage-controlled oscillator,” in Proc. ISSCC, 1999, pp. 402–403. [11] B. D. Leeson, “A simple model of feedback oscillator noise spectrum,” Proc. IEEE, vol. 54, pp. 329–330, Feb. 1966.