HIGH VOLTAGE DC CONVERTER SYSTEMS MODELING, SIMULATION AND ANALYSIS

A thesis submitted in partial fulfillment of the requirements for the degree of Master of Science in Engineering

BY

MANISH DALAL B.E., Gujarat University, India, 1992

2009 Wright State University

WRIGHT STATE UNIVERSITY SCHOOL OF GRADUATE STUDIES

July 7, 2009 I HEREBY RECOMMEND THAT THE THESIS PREPARED UNDER MY SUPERVISION BY Manish A. Dalal ENTITLED High Voltage DC Converter Systems Modeling, Simulation and Analysis BE ACCEPTED IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEGREE OF Master of Science in Engineering.

___________________________ Dr. Marian K. Kazimierczuk Ph.D. Thesis Director

___________________________ Dr. Kefu Xue Ph.D. Department Chair Committee on Final Examination ___________________________ Dr. Marian K. Kazimierczuk Ph.D.

____________________________ Dr. Kuldip S. Rattan Ph.D.

____________________________ Dr. Ray Siferd Ph.D.

____________________________ Dr. Joseph F. Thomas, Jr., Ph.D. Dean, School of Graduate Studies ii

ABSTRACT Dalal, Manish. M.S.E., Department of Electrical Engineering, Wright State University, 2009. High Voltage DC Converter Systems Modeling, Simulation and Analysis.

The thesis provides insights into important modeling techniques to model the converter system, machine design, control and power stages and integration of the various sub systems to simulate the system level performance. This innovative modeling and simulation project is very relevant to optimizing the system performance, designing the sub circuits, components selection, predicting the system stability and impulse responses. The thesis presents modeling and simulation of three different 270VDC converter systems and comparing their performances against each other. The 270VDC converter system accepts either Generator 3-phase AC voltages or fixed three voltage source followed by single or dual converter stages depending on different topologies. The models developed for the high voltage DC systems are optimized to provide robust controls, close loop regulation and transient performance without any algebraic loop by employing valuable modeling techniques. The detailed modeling approach significantly minimizes the development cost by having the model representation of the actual system before the prototype development to ensure ‘first time right’ designs. The system model developed on industry common software platforms establishes the ‘boiler plate’ to allow the new systems to be modeled simply by delta changes on the base systems.

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Contents 1.0 Introduction ..................................................................................................................... 1 2.0 Thesis Problem Statement, Objectives and Approach ............................................ 3 2.1 Problem Statement ...........................................................................................................................3 2.2 Thesis Objectives ................................................................................................................................4 2.3 Thesis Approach .................................................................................................................................4 3.0 DC Converter System Topology Overview ................................................................. 6 4.0 AC Machine Vector control ........................................................................................... 8 5.0 Converter System Topologies..................................................................................... 18 5.1 Topology 1a - 270VDC High Voltage Converter System (115VAC, 3-Φ, 400 Hz input – 270VDC output) ............................................................................................................. 18 5.2 Topology 1b - Inverter (270VDC input - 115VAC, 3Φ, 400 Hz output) ...................... 37 5.3 Topology 2a - 270VDC High Voltage Converter System (Diode Based).................. 45 5.4 Topology 2b - 270VDC High Voltage Converter System (SCR Based)...................... 62 5.5 Topology 3 - 270VDC High Voltage Converter System (VFG input, 270VDC output)............................................................................................................................................... 71 6.0 MEA (More Electric Aircraft) ........................................................................................ 93 6.1 Power Converter Design Optimization .................................................................................. 94 7.0 Conclusions.................................................................................................................... 97 7.1 Recommendation for Future Work ......................................................................................... 99 7.2 Thesis Contribution......................................................................................................................... 99 References..........................................................................................................................101

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List of Figures Figure 4-1 Machine Stator - Rotor Relationship ...............................................................................8 Figure 4-2 Three phase AC Voltage Source Waveform.............................................................. 10 Figure 4-3 Phase A, B and C to α and β............................................................................................. 10 Figure 4-4 Phase a, b, c to α and β to d and q Relationship..................................................... 12 Figure 4-5 Phase a-b-c to α-β to d-q Model.................................................................................... 12 Figure 4-6 α-β to d-q relationship........................................................................................................ 13 Figure 4-7 Machine Stationary and Rotating Phase Relationship......................................... 14 Figure 4-8 Sine Wave Triangle Modulation...................................................................................... 15 Figure 4-9 Space Vector Modulation Switching States .............................................................. 15 Figure 4-10 SVPWM Neutral and Phase Relationship................................................................. 17 Figure 5-1 Topology 1a Simulink Model ............................................................................................ 18 Figure 5-2 Topology 1a Three Phase AC Source Waveform .................................................... 19 Figure 5-3 Topology 1a AC-DC Converter Simulink Model........................................................ 20 Figure 5-4 Topology 1a AC-DC Converter Power Stage............................................................. 21 Figure 5-5 Topology 1a AC-DC Converter Control Stage........................................................... 24 Figure 5-6 Topology 1a Phase Voltages a-b-c, α-β, θ, d-q Waveforms.............................. 25 Figure 5-7 Topology 1a Phase Currents a-b-c, α-β Waveforms ............................................ 26 Figure 5-8 Topology 1a Vα-Vβ, θ, Iα-Iβ Waveforms......................................................................... 27 Figure 5-9 Topology 1a Va-Vb-Vc, Ias-Ibs-Ics, Ia-Ib-Ic Waveforms................................................ 28 Figure 5-10 Topology 1a AC-DC Converter output Voltage & Current................................ 29 Figure 5-11 Topology 1a DC-DC Converter Simulink Model..................................................... 30 Figure 5-11 Topology 1a DC-DC Converter Power Stage.......................................................... 31 v

Figure 5-12 Topology 1a DC Output Voltage Build Up & PWM Gate Pulses ..................... 32 Figure 5-13 Topology 1a Vref, Vo, Control Signal & Load on Waveform............................... 33 Figure 5-14 Topology 1a Main Line Contactor Simulink Model .............................................. 33 Figure 5-15 Topology 1a Vo & Io Waveform ....................................................................................... 34 Figure 5-16 Topology 1a Load Application Simulink Block....................................................... 35 Figure 5-17 Topology 1a Vo Load Transients................................................................................... 35 Figure 5-18 Topology 1a Vo Load Transients................................................................................... 35 Figure 5-19 Topology 1a Two Converter Stages Cascade Technique................................. 36 Figure 5-20 Topology 1b Inverter System Simulink Model........................................................ 37 Figure 5-21 Topology 1b Inverter System Simulink Model........................................................ 38 Figure 5-22 Topology 1b Inverter Power Stage Simulink Model ............................................ 39 Figure 5-23 Topology 1b Inverter Control Stage Simulink Model .......................................... 40 Figure 5-24 Topology 1b Output Va-Vb-Vc, Ia-Ib-Ic Waveforms ................................................ 41 Figure 5-25 Topology 1b Output AC Load on Transient............................................................. 42 Figure 5-26 Topology 1b Output AC Load off Transient ............................................................ 43 Figure 5-27 Topology 1b DC-Link Current........................................................................................ 43 Figure 5-28 Topology 1b FFT of Output AC Voltage..................................................................... 44 Figure 5-29 Topology 2a AC-DC Converter System ..................................................................... 45 Figure 5-30 Topology 2a Converter System Simulink Model ................................................... 46 Figure 5-31 Topology 2a Converter System Power Stage Simulink Model ....................... 47 Figure 5-32 Two Winding Transformer Model................................................................................ 48 Figure 5-33 Transformer Two-Winding Model ............................................................................... 50

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Figure 5-34 Transformer Winding Voltage, Current and Winding Resistance Model Per Phase................................................................................................................ 51 Figure 5-35 18 Pulse Transformer Model Primary and Secondary Phase A ..................... 52 Figure 5-36 18-Pulse Transformer Model 3-Phase Secondary Windings Interconnections ............................................................................................................... 53 Figure 5-37 PLECS Model for 18-Pulse Transformer................................................................... 54 Figure 5-38 DC Converter Model Based on Three Phase 18-Pulse Transformer Model........................................................................................................... 56 Figure 5-39 Transformer Primary and Secondary Voltage Waveforms with 20º Phase Shifted Configuration................................................................................ 57 Figure 5-40 Transformer Primary and Secondary Voltage Waveforms with 20º Phase Shifted Configuration................................................................................ 58 Figure 5-41 Topology 2a Output Voltage Build up Waveform................................................ 59 Figure 5-42 Topology 2a Output Voltage 18-pulse ripple......................................................... 60 Figure 5-43 Topology 2a IPT Leg & Output DC Current.............................................................. 61 Figure 5-45 Topology 2b Converter System Simulink Model ................................................... 63 Figure 5-46 Topology 2b Converter System Power Stage Simulink Model ....................... 64 Figure 5-47 Topology 2b Converter Output Voltage-Current Buil-up.................................. 65 Figure 5-48 Topology 2b Output Voltage 18-pulse ripple......................................................... 66 Figure 5-49 Topology 2b Output Voltage Load on & Load odd Transient ......................... 67 Figure 5-50 Topology 2b Input AC Voltage 18-pulse ripple...................................................... 68 Figure 5-51 Topology 2b IPT Leg & Output DC Current.............................................................. 69 Figure 5-52 Topology 2b SCR Bridge DC Output Voltages........................................................ 70

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Figure 5-53 Topology 3 VFG to AC-DC Converter System......................................................... 71 Figure 5-54 Topology 3 VFG to AC-DC Converter System Simulink Model ........................ 72 Figure 5-55 Topology 3 Power Stage Simulink Model................................................................. 73 Figure 5-56 Topology 3 Synchronous Machine Simulink Model............................................. 73 Figure 5-57 Synchronous Machine Detailed Simulink Model .................................................. 74 Figure 5-58 Exciter Machine and Main Machine Model Parameters ................................... 75 Figure 5-58 Topology 3 Load Application Simulink Model ........................................................ 76 Figure 5-59 Topology 3 High Speed, Output Voltage build-up ............................................... 78 Figure 5-60 Topology 3 High Speed Output voltage Ripple ..................................................... 78 Figure 5-61 Topology 3 High Speed Load-on & Load-off Transient ..................................... 79 Figure 5-62 Topology 3 High Speed VFG AC Voltage Waveform........................................... 80 Figure 5-63 Topology 3 High Speed VFG AC Current Waveform ........................................... 81 Figure 5-64 Topology 3 High Speed Exciter Field Current Waveform ................................. 82 Figure 5-65 Topology 3 High Speed V, I Waveforms ................................................................... 83 Figure 5-66 Topology 3 Low Speed Output voltage Ripple ...................................................... 84 Figure 5-67 Topology 3 Low Speed Load-on & Load-off Transient ...................................... 85 Figure 5-68 Topology 3 High Speed Load-on & Load-off Transient VFG Side ................. 86 Figure 5-69 Topology 3 Low Speed Control Loop Signals......................................................... 87 Figure 5-70 Topology 3 Output Load on Feed-Forward ............................................................ 88 Figure 5-71 Topology 3 Low Speed FFT of Output VDC ............................................................. 90 Figure 5-72 Topology 3 High Speed FFT of Output Voltage VDC ........................................... 91

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List of Tables Table 3-1 Converter/Inverter topologies overview..........................................................................6

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Acknowledgements I am greatly thankful and indebted to Dr. Kazimierczuk for being my advisor for the thesis and for enriching me with valuable insights into power electronics, magnetic design, and converter topologies. I am quite amazed by Dr. Kazimierczuk boundless enthusiasm, passion to inspire students, presentation skills and subject knowledge on each of these courses. I am greatly thankful and appreciative to my organization GE Aviation for support and encouragement to my research and development work. I would like to express my heartfelt thanks to Dr. Abbas, Dr.Hao and Mr. Karipides for their guidance in my research work. Last but not the least; I would like to thank my wife, Shital Dalal for her consistent love, support, encouragement and self-sacrifice for the life we experienced together, both in my good time and not so good time.

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Dedicated to my Family Members and Teachers

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1.0 Introduction

The thesis is divided into seven chapters. Chapter 1 discusses the abstract of the thesis. Chapter 2 discusses the introduction, problem statement, thesis objectives and thesis approach to achieve the objectives. Chapter 3 outlines the converter topologies considered for the thesis and brief description of them. Chapter 4 discusses the vector control theory, abc-αβ-dq transformation, space vector and sine wave modulation, modeling of the vector control to lay the strong theoretical foundation before transitioning to modeling and simulation in following chapters. Chapter 5 comprises of the main body of the thesis which outlines all different converter topologies, modeling, simulation and analysis. Chapter 6 discusses the ever increasing demand for the optimized converter system in MEA (More Electrical Aircraft). Chapter 7 concludes the thesis with conclusions, recommendation of future work and contribution of the thesis. The thesis cuts across various areas of the power system modeling such as d-q transformation technique and modeling, machine design and space vector PWM modeling, converter topology, design and modeling, system simulation for a typical aircraft power system architecture while meeting the power quality requirements for steady state operation, transient operation and distortion requirements. In typical aircraft application, main generator output could be CF (constant Frequency) 400Hz or VF (Variable Frequency) 380-760 Hz. The CF or

1

VF power is generated from main generator which is mounted on AMAD (Airframe Mounted Accessory Drive) on the engine. In some applications, primary power requirement is 270VDC. In such cases, VF or CF output is fed to converter to generate 270VDC or VF or CF with integrated rectifier modules on the flange can provide 270VDC. The thesis probes into various design considerations, modeling and simulation 270VDC system. The primary focus would be on modeling of the power system and predicting the performance. The thesis will cover various topologies, trade study, machine design and optimization, converter architecture, design and optimization and system level performance.

2

2.0 Thesis Problem Statement, Objectives and Approach 2.1 Problem Statement State of art technological advances in area of high efficiency, high density power generators and power converters has contributed significantly to fulfill the power requirements on the aircraft. More and more integrated power solutions are becoming reality to save the weight and volume on aircraft where the weight and space is premium. The tools and techniques to design, model and simulate the power system can play invaluable role in predicting and optimizing the power system performance while meeting the stringent environmental requirements. Aircraft environment is posing significant challenges on designer to optimize the performance while meeting the environment and EMI (Electro Magnetic Immunity) requirements. The typical challenging requirements in designing the power systems are to meet the power quality of MIL-STD-704 specifications, EMI performance requirements per MIL-STD-461E and other environment requirements such as temperature, altitude, vibration, and shock per MIl-STD-810F. Many of the simulation packages are difficult to use, often time consuming and inefficient for simulating dynamics of the power switches. This thesis proposes a Simulink based modeling approach for control stage and PLECS package used to simulate the power stage. PLECS toolbox is integrated in Simulink package to integrate the system level simulation. 3

Various simplified techniques for simulating the control and power stage is demonstrated in the thesis which results into optimized modeling approach with faster computation.

2.2 Thesis Objectives The primary objectives of the thesis are multipronged as outlined below. •

The complete converter system model developed on industry common software platforms to provide the ‘boiler plate’ to allow the new systems to be modeled simply by delta changes on the base systems.

•

Demonstrate detailed system modeling and simulation approach for DC Converter systems, Inverter Systems and AC synchronous machine.

•

Demonstrate transitioning from converter model into inverter model

•

Demonstrate detailed modeling and simulation of the d-q transformation for motor control and how it relates to theory.

•

Demonstrate the modeling and simulation of the complex 18-pulse transformer using Matrix techniques.

•

Electrical Power Quality Analysis such as output voltage regulation, output voltage ripple, transient performance, FFT for the converter systems and compare the performance.

2.3 Thesis Approach The thesis approach is outlined as below. •

Understanding theory of operation for vector control and PWM.

•

Identify three different converter topologies for 270VDC converter system and implement them using the system level modeling. The topologies were 4

selected based on my experience at work and selecting the converter power switches as SCR, IGBT and Diodes to get the good understanding of the power stage modeling. •

Use of Matlab/Simulink for system model design and PLECS for power stage design integrated into Simulink model.

•

Model simulation and detailed analysis for the performance of the various converter systems

•

Identify the key techniques to simulate more effectively such as transformer modeling, linking the multi-stage of the converters, loads application/removal.

•

Thesis conclusion

5

3.0 DC Converter System Topology Overview

For 270VDC high voltage system, three topologies explored in detailed are as illustrated in Table 3-1.

Topology

Input

1a

Three phase voltage source 115, 400 Hz, 3-phase

1b

2a

2b 3

Output (Steady State)

Converter Stage#1

Converter Stage#2

Switching Technology

AC-DC (PFC stage) DC-DC(Isolation)

IGBT

300±5 VDC

270±5 VDC 115VAC, 400 Hz, 3phase

DC-AC (Inverter)

IGBT

Three phase voltage source 115, 400 Hz, 3-phase

270±5 VDC

AC-DC (Isolation)

-

Diode

Three phase voltage source 115, 400 Hz, 3-phase Three phase Variable Frequency Generator

270±5 VDC 270±5 VDC

AC-DC (Isolation)

-

SCR

AC-DC

-

Diode

Table 3-1 Converter/Inverter topologies overview All three topologies provide 270VDC isolated output. Per Topology 1, system receives three phase, 115VAC, and 400 Hz power from source and converts it to 300VDC first using IGBT based PFC converter stage and then converts it to 270VDC using IGBT based isolated DC-DC converter stage. This topology can be modeled as a bidirectional converter where topology 1a is converter and topology 1b is inverter.

6

Per Topology 2, system receives three phase, 115VAC, and 400 Hz power from source and converts it to isolated 270VDC output using diode or SCR based converter stage. Diode based converter will provide unregulated isolated output while SCR based converter stage will provide regulated isolated output. Per topology 3, system front end is 3-phase VFG (variable frequency generator) which generates 3-phase variable AC voltages and variable frequencies. This VFG output is rectified using diode bridge to generate 270VDC output.

7

4.0 AC Machine Vector control

Vector control principle can be implemented on AC machine if the feedback control of the system is performed in the rotating frame. By working out the orthogonal rotating reference frame, all the AC quantities are converted to DC values correspond to the peak value of the AC waveform. The orthogonal relationship between the two currents makes is practical to use two separate control loops to achieve desired

X

bs

cf

a axis

levels of d and q values.

af

X

as

S N

c axis

X cs

bf

b axis

X

Ø

Figure 4-1 Machine Stator - Rotor Relationship Three phase winding along with its stator lamination core is called an armature. Once an armature is connected to a three symmetrical phase power source, it will flow symmetrical currents in the three phases winding. These current will generate the rotating mmf (magneto motive force) in the air gap. 8

Current ia generates pulsating mmf and can not rotate. However, this mmf is a combination of two rotating mmf rotating in opposite direction with the same magnitude. Therefore three phase spatially symmetrical winding with timely symmetrical three phase currents (same magnitude and 120° apart) generates a rotating mmf. Similarly, a two phase spatially symmetrical winding with timely symmetrical two phase currents (same magnitude and 90° apart) also generates a rotating mmf. Therefore, we can use a two phase α, β winding to replace a three a, b, c winding as long as both of them generate the same rotating mmf. This forms the basis of Clarke Transformation. Phases a, b and c are stationary vectors which are phase shifted by 120º from each other. α and β are perpendicular stationary vectors representing the phases a, b and c in stationary frame. As shown in Fig 4-3, at wt=0, a=0,b=-0.866, c=0.866 resulting into α=0 and β=1. At wt=30º, α=0.5 and β=-0.866. The green vector rotates counterclockwise and completes one rotation of -п/2 to 3п/2 over one cycle of the phases a, b and c. The projections of rotating green vector on α and β axis change continuously over a period of one cycle of 360º.

9

Phase a, b and c Unit based, 400 Hz

1

a

0.8

b

c

0.6

0.4

Volts

0.2

0

-0.2

-0.4

-0.6

-0.8

-1

0

0.001

0.002

0.003

0.004

0.005 Time (seconds)

0.006

0.007

0.008

0.009

0.01

Figure 4-2 Three phase AC Voltage Source Waveform

ß

a=0 ß=1

a

ß a=0.5 ß=-0.866

a

ß a=1 ß=0

a

Figure 4-3 Phase A, B and C to α and β 10

As shown in Figure – 4-3, phase a, b and c can be represented in terms of stationary vectors α and β. α is aligned with phase a. α is effectively phase a, b and c projected on a-axis (same α axis). The β vector is perpendicular to α vector. The β can be derived by reflecting phases a, b and c on β axis. As shown in Figure-Y, stationary vectors α and β will take values from 0 to 1 over a complete cycle of 360º. The conversion matrix of a-b-c to α-β can be represented as,

2 [ia + cos 120°ib + cos 240°ic ] 3 2 iβ = [ 0 + sin 120°ib + sin 240°ic ] 3

iα =

This will result into,

1  1 − iα  2  2 i  =  3  β  3 0  2

1  i  a 2  i   3  b  − i  2   c  −

Now as ia + ib + ic =0, above matrix can be simplified as,

iα   1 i  =  1  β   3 This is called Clarke transformation.

11

0  i  2  a  i 3   b 

937 VFG/Rectifier - 26250 RPM - 75 kW transient POR Voltage 1 a

b

c

Volts

0.5

0

-0.5

-1 0

0.001

0.002

0.003

0.004

0.005 Time (seconds)

0.006

0.007

0.008

0.009

0.01

0.006

0.007

0.008

0.009

0.01

0.006

0.007

0.008

0.009

0.01

α and β 1

α

β

Amperes

0.5

0

-0.5

-1 0

0.001

0.002

0.003

0.004

0.005 Time (seconds) Rotating Vector

Angle (-π to + π )

4

2

0

-2

-4

0

0.001

0.002

0.003

0.004

0.005 Time (seconds)

Figure 4-4 Phase a, b, c to α and β to d and q Relationship

Sine Wave 1 Sine Wave 1

Gain

3ph ->SRF1

Scope 7

Sine Wave 2

simout To Workspace

atan 2 Trigonometric Function 1

Figure 4-5 Phase a-b-c to α-β to d-q Model

12

The stationary frame α-β can be converted to the rotating frame d-q if the reference angle θ is known. This is also called rotor position angle which is the angle between α-axis and d-axis. Therefore, if the d-axis is aligned with α-axis then the reference angle θ is zero. The d-q frame rotates depending on the values of α and β.

q axis

β axis

d axis

θ

a axis

α axis

Figure 4-6 α-β to d-q relationship id is a projection of α and β on d-axis. iq is a projection of α and β on q-axis. This results into Park transformation,

id   cosθ i  =   q  − sin θ

sin θ  iα    cosθ  i β  .

Two separate control loops to achieve the desired values of d and q can be optimized for the vector control of the AC machine. To implement the Vector Control principles, we convert the stationary phases a, b, and c windings to stationary frame α and β to rotating frame d and q windings that rotate at the

13

same speed as the rotor by Park transformation and control the d and q currents in the new windings accordingly. β axis

b axis

ω d axis

q axis

ω

θ ω

a axis

α axis

c axis

Figure 4-7 Machine Stationary and Rotating Phase Relationship Triangular Sine Wave vs. Space Vector PWM control: Over last few years, many different PWM (Pulse Width Modulation) techniques have been developed and studied to gain following primary objectives: Wider linear modulation range; improved switching performance with lower switching losses; less THD Total harmonic Distortion); Simple implementation and less computation time. Figure 4-8 shows the triangular sine wave modulation, where the carrier signal frequency is usually at least 20X the modulation frequency. The PWM has two modes of operation. 1) Linear Mode – In Linear mode, the modulation signal peak is less than or equal to the carrier signal peak. 2) Nonlinear Mode – In Nonlinear mode, the modulation signal peak is above the carrier signal peak which would cause THD of the output waveform to increase. 14

Ia

Vdc

Ib

Va

Vb

Vc

Ic

Vn

Figure 4-8 Sine Wave Triangle Modulation In a linear modulation range, the line-to-neutral peak voltage would be

VLN (max) = 0.5Vdc Figure 4-9 shows the switching states of the space vector PWM. b r

V3 (010)

r V4 (011)

r V7 (111) r V8 (000)

β

r V2 (110)

r Vr

r

r T1V1 / Tz

c V (001) 5

r V (100) T2V2 / Tz 1

α,a

V6 (101)

Figure 4-9 Space Vector Modulation Switching States The reference vector Vr rotates along inside the hexagon. The hexagon has six sectors representing one sampling interval and eight switching states. Sector 1 represents the sine wave from п/2 to 5п/6; sector 2 represents the sine wave from

15

5п/6 to 7п/6 and so on. The V1 through V6 are active vectors. The V0 and V7 are zero vectors. For linear modulation range, maximum line-to-neutral voltage peak would be,

2 VLN (max) = Vdc * cos 30 3 VLN (max) = 0.577Vdc Therefore, space vector modulation provides 15% (0.577/0.5) better dc bus voltage utilization compared to triangular sine wave modulation. Another important aspect of the space vector modulation is that for sinusoidal waveforms, the reference vector Vr must inscribe the circle. However for non-sinusoidal applications in some cases for motor applications, where noise in ac waveforms is filtered by the motor inductances, the reference vector is extended to the periphery of the hexagon. Space Vector limitations: The space vector modulation can not be used for inverter/converter applications which have separate neutral. This is because space vector PWM generates zero sequence voltage which is third harmonics. As shown in Figure A, the zero sequence voltage exists for space vector modulation. However if the neutral is not used then the effective inverter three phase voltages would be perfect sine wave signals neglecting the higher order harmonics.

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Vα, Vβ

Va, Vb and Vc with respect to DC-Link common

(Va+Vb+Vc )/3 Zero sequence (Third Harmonics)

(Va)- (Va+Vb+Vc )/3, (Vb)- (Va+Vb+Vc )/3, (Vc)- (Va+Vb+Vc )/3

Figure 4-10 SVPWM Neutral and Phase Relationship

17

5.0 Converter System Topologies 5.1 Topology 1a - 270VDC High Voltage Converter System (115VAC, 3Φ, 400 Hz input – 270VDC output) This topology consists of two stages of the converters. 1. AC-DC PFC Converter 2. DC-DC Isolated Converter The front end of the system is power factor corrected AC-DC converter which accepts 115Vac, 400 Hz, 3-phase voltages and convert it to 300 VDC. The second stage of this system is DC-DC converter which accepts 300 VDC and converts it to regulated 270VDC output. This system provides isolated output from input. System Modeling: The system model is developed on Simulink platform. The power stage is developed on PLECS platform and integrated into Simulink. The control stage is developed in Simulink. The system model consists of the multi stage sub systems. 301.2

270

VDC Link

VDC Output

Scope2

Phase A Vdc VAC IN

VDC_Link M

VDC Link S

Phase B

Idc IDC_Link S

IDC_Link M

Scope1

I_Load IDC Link M

Phase C

DC-DC Converter

AC-DC Converter

Load

Load (Amps)

Figure 5-1 Topology 1a Simulink Model

18

AC Source For this system, three phase fixed voltage source is used. The waveforms for the 3phase voltages are illustrated as below.

Figure 5-2 Topology 1a Three Phase AC Source Waveform

19

AC-DC power converter with PFC

Out1

In1

SS_abc Vs_qd* Stationary

Space Vector Sequencer

Gate Drive

1 VDC_Link M 2 IDC_Link M 1 VAC IN

-KGain

v_dc AC_IN

iaigbt i_dcbus

DCLOAD

Scope

i_ain

PLECS Circuit

Iabco2

Inverter 3ph->SRF2Output Current

VAC IN Icap

Gate_abc1

VAC IN1

Power Stage

Inverter Output Current

Scope1 2 IDC_Link S

3ph->SRF1

atan2

In1Out1

Trigonometric Function1

Mod

In2 In4 In1

Out1

In3 In5

Control Stage

SRF->RRF2

Scope2

Figure 5-3 Topology 1a AC-DC Converter Simulink Model

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Inverter Output Voltage

Power Stage

Figure 5-4 Topology 1a AC-DC Converter Power Stage The power stage of AC-DC converter is modeled on PLECS platform. The input 3phase voltages pass through the LC filter circuit and then applied to 3-phase IGBT bridge. The gates of the IGBTs are controlled through SVPWM techniques to ensure that the 3-phase AC source current is in phase with the 3-phase AC source voltages by cancelling the leading AC current component caused by the LC filter in the front end. Control Stage In power converter applications, it is important to maintain the power factor of the source AC voltages close to unity. For this, close loop control with d-q transformation approach becomes integral to the feedback control system. The feedback control system is performed in rotating reference frame that is synchronized to an angle of 21

the incoming 3-phase voltages. Three phase AC voltages Va, Vb and Vc are converted to Vα and Vβ. The reference phase angle θ is measured by estimating arctangent of (Vb/ Va). The objective of achieving the close to unit power factor can be realized by ensuring that three phase source currents Ia, Ib and Ic are in phase with Va, Vb and Vc. The reference phase angle θ is the common parameter to tie voltage and current phase relationship. Therefore, the three phase measured currents Ia, Ib and Ic is converted to Iα and Iβ. In order to achieve unity power factor, Vα should be in phase with Iα. In order to achieve this, measured stationary Iα and Iβ are converted to rotating frame Id and Iq using the reference phase angle of θ. Therefore all AC quantities of Iα and Iβ are converted to DC quantities Id and Iq. This DC levels are corresponding to the peak values of AC waveform of AC currents. The reference angle θ is at zero position when reference vector is lined up with α-axis. At θ=0, both α-axis and d-axis are in phase. Therefore, id is the component of the current which is in phase with source AC voltage and iq is the component of the current which is 90º out of phase with the source AC voltage. The orthogonal relationship between id and iq makes it practical to use two separate PI control loops to adjust the output voltage of the inverter to force id and iq to follow a desired command. So, id represents the current which is phase with source voltage. This is the parameter which controls real power transferred between the AC and DC buses. The commanded value of id depends on the third PI control loop that adjusts the current set point id based on the difference between DC output voltage measured across the output filter capacitor and the desired DC output voltage. The voltage feedback loop will force the in-phase current (id) to be at the proper level to keep the DC bus of the inverter at a constant

22

output voltage under the influence of varying DC load currents. Importantly, the command value of iq can be set to a constant value in order to inject a lagging reactive current in the line which cancels the leading line currents being generated by the input LC EMI filter.

23

id*

Ki

1/s

Gain1

Integrator

Kp i_switch

2 In4

Gain3

Id PI Loop -1

SRF->RRF2 Stationary to Rotating Transformation

Ki

iq*

Gain7

1/s

-1 RRF->SRF

Gain4 Integrator1 Rotating to Stationary Transformation

Kp Scope3

Gain5

Iq PI Loop

8.2 Iq Setpoint To Cancel Capacitive Input Filter -K3 In1

Id=(2/3)(Vo*Io/Vd)

i_inductor

3

Gain6

2/3

1 s

Gain9

Integrator2

Gain8

300 DC Bus PI Loop

DC Bus Regulation Setpoint 4 In3

Divide

1 In2 5 In5

Scope

Divide1

1 Gain10

Scope1

Figure 5-5 Topology 1a AC-DC Converter Control Stage

24

Gain2

1 Out1

The outer control loop is voltage control loop. The DC voltage reference is set to 300 VDC and the feedback is measured from the DC bus. The error between the reference and feedback is fed through the error amplifier to set the inductor current set point iL. The current through the inductor is same as id* - id. So id*=id+iL. The inner loop is the current loop which sets the duty cycle to control the gates of the IGBT to close the loop.

200

Volts

100

Va

Vb

Vc

0 -100 -200 0.005

0.01

0.015

0.02

0.015

0.02

0.015

0.02

0.015

0.02

Time (seconds) 200

Volts

100

α

β

0 -100 -200 0.005

0.01 Time (seconds)

Phase (-π to + π )

4 2

θ

0 -2 -4 0.005

0.01 Time (seconds)

200

Vd,Vq

150

Vd

100 50 Vq 0 -50 0.005

0.01 Time (seconds)

Figure 5-6 Topology 1a Phase Voltages a-b-c, α-β, θ, d-q Waveforms

25

200 150 Va

Inverter Cirrent

100

Vb

Vc

50 0 -50 -100 -150 -200 0.005

0.01

0.015

0.02

0.015

0.02

Time (seconds)

200 150 Iα

100

Iβ

Iα ,Iβ

50 0 -50 -100 -150 -200 0.005

0.01 Time (seconds)

Figure 5-7 Topology 1a Phase Currents a-b-c, α-β Waveforms

26

200

Inverter Cirrent

100

Vα

Vβ

0

-100

-200 0.005

0.01

0.015

0.02

0.015

0.02

0.015

0.02

Time (seconds)

4

θ (-π to π )

2

θ 0

-2

X: 0.005002 Y: -1.566

-4 0.005

0.01 Time (seconds)

200

Inverter Cirrent

100

Iα

Iβ

0

-100

-200 0.005

0.01 Time (seconds)

Figure 5-8 Topology 1a Vα-Vβ, θ, Iα-Iβ Waveforms

27

200

Volts

100

Va

Vb

Vc

0

-100

-200 0.005

0.01

0.015

0.02

0.015

0.02

0.015

0.02

Time (seconds)

Inverter Current

200

100

Ia

Ib

Ic

0

-100

-200 0.005

0.01 Time (seconds)

200

Input Current

100

Ia

Ib

Ic

0

-100

-200 0.005

0.01 Time (seconds)

Figure 5-9 Topology 1a Va-Vb-Vc, Ias-Ibs-Ics, Ia-Ib-Ic Waveforms

28

Output Voltage and Current

VDC = 300VDC

IDC at 40kW Load

Figure 5-10 Topology 1a AC-DC Converter output Voltage & Current

29

DC-DC Isolated Converter 3

1

IDC Link M

Vdc 2 Idc

Vo

Gate_abc1

Io

1

PLECS Circuit

In1

VDC Link S

Zero-Order Hold1

Scope3 Scope2

Idcl

Zero-Order Hold6 ILoad

Iload

>0

Vo

Zero-Order Hold

Compare To Constant

D= (VO+VL)/VDCL

1

Power Stage

Gain6

Scope Saturation

2 VL

I_Load

In1Out1 I_Load

1

In1 Out1

PI Loop 2

I_Load_SP

scale V_POR_Sense

-1 Iload

Zero-Order Hold3

Repeating Sequence

PI Loop 1

Main Line Contactor Logic

Vref 270VDC

Scope1

Zero-Order Hold7

feed forward control Iload_before_cap

Zero-Order Hold5

Figure 5-11 Topology 1a DC-DC Converter Simulink Model

30

Gain

Power Stage

Figure 5-11 Topology 1a DC-DC Converter Power Stage The power stage receives input DC link voltage from first AC-DC converter output. The power stage consists of front end H bridge IGBT converter. The gates of these IGBTs are controlled by the control stage sine wave triangle PI controller. The output of the H bridge converter is fed to the high frequency isolation transformer. The parameters of the transformer are as follows. L1=10mH, L2=10mH, M=9.9999mH. Therefore the co-efficient of coupling K=

M =0.99999 and leakage inductance = Lleakage=L1 *(1- k 2 ) = 0.2uH. The L1 * L 2

transformer secondary is connected to full wave rectifier bridge to converter to DC output. The output filter stage consists of the LC filter. The filter inductor is 56uH and filter capacitor is 100uF which gives a filter cut off frequency of ~2125 Hz. Control Stage The control stage consists of the outer voltage loop which receives 270VDC reference compares to the output voltage feedback to generate the error signal. The error signal passes through PI error amplifier and creates the DC load current set point. The

31

feed forward control is used by using the summing circuit to add the difference between the measured load current and load current before the filter capacitor to provide the phase boost. The summing circuit output passes through another error amplifier and generates the inductor voltage set point VL. The simplified form of the relationship between the duty cycle and VL is given by VDCLink* D = Vo + VL which results into D=

Vo + VL . In order to match the sine wave triangle PWM logic the duty cycle VDCLink

reference is adjusted by subtracting it from 1. The next stage compares the triangle waveform to the control signal and generates the gate pulses to drive the IGBTs.

PWM Gate Pulses

DC Output Voltage Build up

Figure 5-12 Topology 1a DC Output Voltage Build Up & PWM Gate Pulses

32

Vref = 0 to 270VDC

Vout = 0 to 270VDC

Vcontrol

Iload (10 amp to 150 amps)

Figure 5-13 Topology 1a Vref, Vo, Control Signal & Load on Waveform MLC (Main Line Contactor) Logic 1 I_Load

1

Switch1

I_Load_SP 2

V_POR_Sense

10 Pre Load MLC Delay

Q

S

!Q

R

boolean

> 265

Data Type Conversion

Compare To Constant1

S-R Flip-Flop

Figure 5-14 Topology 1a Main Line Contactor Simulink Model

33

Main line contactor is located between the three phase AC voltage source and converter input. The control logic measures the output DC voltage and ensures that the contactor closes when output voltage exceeds 265 VDC. When MLC closes the loads on the bus gets applied to the converter. System performance Steady State Performance

Voutput

Ioutput

Figure 5-15 Topology 1a Vo & Io Waveform Transient performance Load application and removal is implemented by model developed as below. 150 Constant4

Repeating Sequence4 1 Load

Repeating Sequence3

Switch1

10 Constant3

34

Selector

Figure 5-16 Topology 1a Load Application Simulink Block

Load off

Load on

Output Load 10 amps to 150 amps

Figure 5-17 Topology 1a Vo Load Transients

Load on

Load off

Output Load 10 amps to 150 amps

Figure 5-18 Topology 1a Vo Load Transients

35

Power Quality Both transient and steady state performance meets the MIl-STD-704F power quality. The loads on transient results into output voltage dip from 270VDC to 256 VDC which is above minimum required MIl-STD-704F acceptable limit of 200VDC. The loads off transient results into output voltage overshoot from 270VDC to 284 VDC which is below maximum required MIl-STD-704F acceptable limit of 330VDC. The steady state voltage regulation for 270VDC system per MIl-STD-704F specifications is < 6 Vrms. The steady state ripple voltage observed for this system is SRF3

i_dcbus Gate_abc1

POWER STAGE

Scope1

Transfer Fcn

iaigbt

DC Bus

SRF->RRF3

i_ain Iabco2 2

Icap

Vabc I_Load_S

2 Power(kW)

Iload

Iabc

VAC Out

Pnom

3ph->SRF2

1 Vabc

Subsystem

Power Stage

Phase Idq_sw

SRF->RRF1

Out1

Vdq Idq_op

3ph->SRF1 In1 Out1

In1

SRF->RRF2

Out1

Clock1 Angle

Mod

Figure 5-21 Topology 1b Inverter System Simulink Model

38

Control

Inverter Output Voltage

Power Stage The power stage of DC-AC inverter is modeled on PLECS platform.

Figure 5-22 Topology 1b Inverter Power Stage Simulink Model The input DC voltage is applied to 3-phase IGBT bridge. The gates of the IGBTs are controlled through SVPWM techniques to ensure that the 3-phase AC output current is in phase with the 3-phase AC source voltages by cancelling the leading AC current component caused by the LC filter in the front end.

39

Control Stage 2 Idq_sw Id

Vd

200

3 Vdq

1/s

Gain12 Repeating Sequence

Ki

Integrator3

2

Kp

Gain13

Gain16 Id PI Loop

4 Idq_op

Vq

200

-1 RRF->SRF

1/s

Gain10 2

1 Out1

Gain2

Integrator4

Ki

0 Iq Setpoint

1/s

Gain15 Integrator5

1/s

Gain1 Integrator1

Iq

Gain14

Kp Gain3 Id PI Loop

1 Phase

Figure 5-23 Topology 1b Inverter Control Stage Simulink Model Close loop control with d-q transformation approach becomes integral to the feedback control system. The feedback control system is performed in rotating reference frame that is synchronized to a reference angle of the output 3-phase voltages. Measured three phase AC voltages Va, Vb and Vc are converted to Vα and Vβ. The reference phase angle θ is set for -∏ to +∏. Vd and Vq are generated using the reference angle θ and Vα and Vβ. Vd reference voltage of 162 (115*1.4142) and Vq reference voltage of 0 is set for close loop control. The outer voltage PI control loop receives Vd and Vq references and feedback and provides Id and Iq references for the inner PI control loop. The reference phase angle θ is the common parameter to tie voltage and current phase relationship. Therefore, the three phase measured currents Ia, Ib and Ic is converted to Iα and Iβ which are again converted into rotating frame Id and Iq using the reference phase angle of θ. The inner PI current loop uses the feed-forward control for improving the transient response. Close to unity power 40

factor can be realized by ensuring that three phase currents Ia, Ib and Ic are in phase with Va, Vb and Vc.

AC 3-Phase Voltage

AC 3-Phase Current

Figure 5-24 Topology 1b Output Va-Vb-Vc, Ia-Ib-Ic Waveforms

41

AC 3-Phase Voltage

AC 3-Phase Current

Load On 1kW-40kW

Figure 5-25 Topology 1b Output AC Load on Transient

AC 3-Phase Voltage

AC 3-Phase Current

Load Off 40kW-1kW

42

Figure 5-26 Topology 1b Output AC Load off Transient

DC Link Input Voltage

DC Input Current Output Load On 1kW-40kW

Figure 5-27 Topology 1b DC-Link Current

43

180 160

Vabc FFT Fundamental 400 Hz, 162 Vpeak (115Vrms) at 40kW load

140

Magnitude (V)

120 100 80 60 40 20 0

0

0.2

0.4

0.6

0.8 1 1.2 Frequency (Hz)

1.4

1.6

Figure 5-28 Topology 1b FFT of Output AC Voltage

44

1.8

2 4

x 10

5.3 Topology 2a - 270VDC High Voltage Converter System (Diode Based) This topology consists of single stage of the converter. The front end of the system is three phase voltage source which is applied to input inductor which simulates the machine leakage inductance if machine is used instead of fixed voltage source. This is followed by 18-pulse isolation transformer which is followed by the Diode or SCR based converter. The output of the SCR converter is fed through the IPT (Inter phase Transformer) circuit to generate 270VDC. This output passes through the output LC filter circuit to provide filtered 270VDC output. This system provides isolated output from input. Interphase Transformer

Secondary

+VDC

Primary

Return

Figure 5-29 Topology 2a AC-DC Converter System

45

System Model

L1 M12 M13 M14 M15 M16 M17

M12 L2 M23 M24 M25 M26 M27

M13 M23 L3 M34 M35 M36 M37

M14 M24 M34 L4 M45 M46 M47

M15 M25 M35 M45 L5 M56 M57

M16 M26 M36 M46 M56 L6 M67

M17 M27 M37 M47 M57 M67 L7

General Inverse Matrix Multiply

(LU) LU Inverse

Matrix Multiply Matrix Multiply

Leg A Matrix Leg A Voltages

1 s

Vb Vc Vout

Leg B Currents Ib'

Out2

Integrator

Matrix Multiply1

Leg c Voltages 270.6 Display

Out3

POWER Circuit

Leg c Currents 1 s

Matrix Multiply

Leg B Voltages Ia'

Integrator2

1 s

Matrix Multiply3

Va

Leg A Currents

Scope6

Scope2

Out4

Ic'

Scope3

Out5

Integrator1

ic1

Scope4

ic2

Scope1 105.5

Vrms

v abc

Vabc

Scope5

ic3

Constant

Iout

VSCF Control

44.55 Display2

Power Circuit

Figure 5-30 Topology 2a Converter System Simulink Model

46

Power Circuit

Figure 5-31 Topology 2a Converter System Power Stage Simulink Model

47

Transformer Model Transformer Two-Winding Matrix Model The two-winding transformer circuit can be shown as Figure 10, where Lp = Primary leakage inductance Lm = Primary magnetizing inductance Ls = Secondary leakage inductance M12 = Mutual Inductance between primary and secondary

i1

M12

Lp

Ls

i2

Np : Ns Lm

v1

Ideal Transformer

v2

Figure 5-32 Two Winding Transformer Model

This model can be represented in matrix form as follows. V1   L1 V  = s   M 12  2 

M 12  i1  L2  i2 

Transformer Multi-Winding Matrix Model For multi-winding transformer, the coupling between the primary-secondary and secondary-secondary results into complex matrix. The mutual inductance between

48

all the windings needs to be incorporated in the matrix. This would result in matrix as below. V1   L1 V  M  2  21 •  = s  •    •   • Vn   M n1  

M 12 L2 • • M n2

• • M 1n  i1  • • M 2 n  i2  • • •  •    • • •  •  • • Ln  in 

Where Ln = Self inductance of nth winding Mnm = Mutual inductance between nth winding and mth winding Vn = voltage across nth winding in = current through nth winding The above matrix is symmetric and the all the parameters of this matrix can be calculated or measured. In order to implement this model in simulation, the derivative must be removed and replaced with the integral. This can be achieved by inverting the matrix as below. So that the model becomes the transconductance devices such that the currents are controlled through the integral of a linear combination of the voltages. i1   L1 i    2  1  M 21 •  =  •   s •   • in   M n1  

M 12 L2 • • M n2

• • M 1n  • • M 2 n  • • •   • • •  • • Ln 

−1

V1  V   2 •    •  Vn   

49

Model Implementation The two winding model can be implemented as shown in Figure 11. It shows that the inductance matrix is multiplied to the voltage and then integrated to provide the current control. This model can be extended to any number of windings. i1

i2 R1

+

R2

+

v1

v2

-

v1

v2

i1

Mux

Inductance Matrix

Matrix Inverse

Matrix Multiply

i2 DeMux

1/s

Figure 5-33 Transformer Two-Winding Model The transformer design is based on 7-winding configuration where w1 is primary, w2, w3 and w4 are identical three separate main secondary windings, w5, w6 and w7 are zigzag secondary windings to achieve 18-pulse configuration.

50

Figure 5-34 Transformer Winding Voltage, Current and Winding Resistance Model Per Phase The Figure 14 circuits are integrated into sub circuit per Figure 15.

51

Figure 5-35 18 Pulse Transformer Model Primary and Secondary Phase A Figure 16 shows the three phases of the transformer and 7 windings per phase. The primary windings w1 are connected in delta configuration as shown. Secondary windings w2 through w6 are connected to achieve transformer configuration per Figure 13. The 9-phases are generated from this configuration which is similar to 20º phase shifted configuration similar to Figure 4. The winding voltages are multiplexed and the winding currents are de-multiplexed as shown. Figure 16 circuits are integrated into sub circuit per Figure 17 which shows the power stage of the DC converter. The transformer power stage as shown in Figure 17 is integrated into DC Converter system model Figure 18.

52

Figure 5-36 18-Pulse Transformer Model 3-Phase Secondary Windings Interconnections

53

Va Ia'

Vb

Vc

Vo Ib' Vi 2

Vi 3

Ic'

TRANSFORMER Circuit

Out4

Vi 1

ic1 Vabc ic2

ic3

Out6

Iipt

Vipt

Circuit 1

Figure 5-37 PLECS Model for 18-Pulse Transformer Figure 19 shows the complete system model with the inductance matrix. This is an extension of Figure 11 for 3-phase multi winding transformer model. The inductance matrix is based on calculating the self and mutual inductances as explained earlier. The winding resistances are calculated as explained earlier and the values are

54

incorporated in the model. The identical inductance matrix is used for all three phases due to symmetry. Figure 20 and 21 depicts the verification the transformer model.

55

Transformer Model Verification

L1 M12 M13 M14 M15 M16 M17

M12 L2 M23 M24 M25 M26 M27

M13 M23 L3 M34 M35 M36 M37

M14 M24 M34 L4 M45 M46 M47

M15 M25 M35 M45 L5 M56 M57

M16 M26 M36 M46 M56 L6 M67

M17 M27 M37 M47 M57 M67 L7

General Inverse

Leg A B and C Matrix Leg A Currents

(LU)

Matrix Multiply

LU Inverse

Matrix Multiply

Leg A Voltages Va

Matrix Multiply Matrix Multiply 1

1 s

Leg B Voltages

Ia '

Vb

Integrator 2

Leg c Voltages Vc

Leg B Currents Vo

1 s

Vi 2

Vi 3

Leg c Currents 1 s

115

Matrix Multiply 2

Ib '

Integrator

Constant

Matrix Multiply

Ic '

TRANSFORMER Circuit

Out4

Integrator 1 Vi 1

Display 1 vabc

Vrms

ic1

Vabc ic2

AC Control ic3

Out6

Iipt

Vipt

Circuit 1

Figure 5-38 DC Converter Model Based on Three Phase 18-Pulse Transformer Model 56

Figure 5-39 Transformer Primary and Secondary Voltage Waveforms with 20º Phase Shifted Configuration

57

300

200

100

0 0.017

0.0195

0.022

-100

-200

-300 Phase 4

Phase 1

Phase 7

Phase 2

Phase 5

Phase 8

Phase 3

Phase 6

Phase 9

Primary AB

Figure 5-40 Transformer Primary and Secondary Voltage Waveforms with 20º Phase Shifted Configuration

58

System Performance Output Voltage Build-up

Output Voltage Build Up

Figure 5-41 Topology 2a Output Voltage Build up Waveform 59

Output Voltage and Current Ripple

18-pulse ripple (over 2.5ms) Steady State Output Voltage

Steady State Output Current

Figure 5-42 Topology 2a Output Voltage 18-pulse ripple 60

Output DC Current and IPT Currents

IPT Leg A, B and C Currents

Steady State Output Current

Figure 5-43 Topology 2a IPT Leg & Output DC Current

61

5.4 Topology 2b - 270VDC High Voltage Converter System (SCR Based) Topology 2b is similar to 2a except the Diodes are replaced with SCR switches and control circuits for the SCR to regulate the output voltages to 270VDC. Interphase Transformer

Secondary

+VDC

Primary

Return

Control Stage

Figure 5-44 Topology 2b AC-DC Converter System

62

Simulation Model:

L1 M12 M13 M14 M15 M16 M17

M12 L2 M23 M24 M25 M26 M27

M13 M23 L3 M34 M35 M36 M37

M14 M24 M34 L4 M45 M46 M47

M15 M25 M35 M45 L5 M56 M57

M16 M26 M36 M46 M56 L6 M67

M17 M27 M37 M47 M57 M67 L7

Matrix Multiply

General Inverse

Matrix Multiply (LU) Matrix Multiply

LU Inverse

Matrix Multiply4 Leg A Matrix 1 s

Leg A Currents

Matrix Multiply Leg A Voltages

Ia'

Leg B Voltages

Integrator2 1 s Scope4

Leg B Currents

115

Vb

Leg c Voltages

Ib'

v abc

270.2

Vo Ic'

Display

Vi2

Integrator1 Leg c Currents Vrms

Scope6

Vc

Integrator 1 s

Constant1

Matrix Multiply1

Va

Vi3 Vabc

115

Out4

Scope2

Switch Constant

VSCF Control

Vi1

POWER Circuit

Iipt

MOD ON

MOD_ON

MOD_A

Scope3

ic1

0

ic2

Gates_A ic3

Display1 TRU Control

Scope5

Out6 Gates_B Vabc

Vipt

MOD_B

Scope8 Ia

1000

Ib

s+1000

Gates_C Vc

Scope1

MOD_C

42.84 Display2

Transfer Fcn

Ic Load

Fire/Blank and MOD1 Circuit1 Vcontrol

Vterm

a b b

b a b

General Inverse

b b a

(LU) LU Inverse1

ipt Matrix

vo

1 s Integrator3

Load Current Load

12

Matrix Multiply 1/(N^2) Matrix Multiply3 Gain Scope7

Load in kW

Figure 5-45 Topology 2b Converter System Simulink Model

63

Scope9

Power Stage:

Figure 5-46 Topology 2b Converter System Power Stage Simulink Model

64

System Performance: Voltage Build Up

Output Voltage Build Up

Output Current Build Up

Figure 5-47 Topology 2b Converter Output Voltage-Current Buil-up

65

Steady state voltage and current

18-pulse output voltage ripple (over 2.5ms) Steady state Output voltage ripple

Steady state Output Current

Figure 5-48 Topology 2b Output Voltage 18-pulse ripple

66

Load on and Load off transients

Load Off - Output Voltage Overshoot

Load On - Output Voltage Dip

Load Off

Load On

Figure 5-49 Topology 2b Output Voltage Load on & Load odd Transient

67

Input AC Current 18-pulse 3-phase

Figure 5-50 Topology 2b Input AC Voltage 18-pulse ripple

68

Output DC Current and IPT Currents

IPT Leg A, B and C Currents

Steady State Output Current

Figure 5-51 Topology 2b IPT Leg & Output DC Current

69

SCR Bridge 1, 2 and 3 Voltages

SCR Bridge#1 DC Output

SCR Bridge#2 DC Output (20º Lead)

SCR Bridge#3 DC Output (20º Lag)

Figure 5-52 Topology 2b SCR Bridge DC Output Voltages

70

5.5 Topology 3 - 270VDC High Voltage Converter System (VFG input, 270VDC output) This topology consists of 3-phase VFG (Variable Frequency Generator) as a source followed by 6-pulse full wave rectifier-bridge to generate 270VDC output. The rectifier bridge and output filter circuit can be integrated on the flange of the generator to provide space saving integrated package design. Alternatively the rectifier bridge and output filter circuit can be a split box located in environmentally controlled area away from the VFG. This topology provides 6-pulse output from the rectifier bridge which is filtered through output filter capacitor to provide clean 270VDC. The close loop control is provided by controlling the field exciter voltage of the machine to regulate the stator three phase AC voltages and thereby regulating the 270VDC output. This system provides isolated output from input.

GVR and POWER Supply Assembly

EXCITER STATOR

N PMG ROTOR

PMG STATOR

S

MAIN ROTOR

PMG STAGE

3-Ø MAIN STATOR MAIN MACHINE

EXCITER ROTOR EXCITER

RECTIFIER ASSEMBLY

VFG (Variable Frequency Generator)

Figure 5-53 Topology 3 VFG to AC-DC Converter System

71

System Model

Iabc

Scope5 26250

pi/30

Wm Vabc

Speed (rpm) Scope6 m_m

m_e

In1 Out1 In1 Out1

Vref 270VDC

Zero-Order Hold7

POWER STAGE m2 Circuit

Vex

PI Loop 1 PI Loop 2 Iex

Scope2 270.2 Vo

Display

ILoad_M DC Load

Iload

Amps

Scope1 10.27 Display2

IL_M

VFG & Rectifier PLECS MODEL

Zero-Order Hold1 Zero-Order Hold2

Figure 5-54 Topology 3 VFG to AC-DC Converter System Simulink Model The system model consists of the power stage and the control circuit integrated to regulate 270VDC output. The VFG can be run at any speed in the range of 14000 rpm to 26000 rpm.

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Power Circuit:

Figure 5-55 Topology 3 Power Stage Simulink Model The power stage consists of the VFG synchronous machine whose 3-phase output is fed to rectifier-bridge to convert to DC. The output filter consists of LC circuit for which filter capacitor is selected at 300uF and filter inductor selection depends on the leakage inductance provided by the Main VFG. If the VFG leakage inductance is sufficient then external filter inductance L is not required. VFG:

Figure 5-56 Topology 3 Synchronous Machine Simulink Model

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The VFG model receives two inputs. •

Speed Wm (rad/s)

•

Excitation voltage Vex (VDC)

The VFG is a synchronous generator which consists of two machines inside. •

Exciter Synchronous Machine

•

Main Synchronous Machine

The exciter machine receives DC excitation voltage on exciter stator. This generates three phase AC voltages on exciter rotor which is rectified using rotating rectifier bridge in exciter rotor. This rectified DC voltage from exciter rotor is fed to the main machine rotor. This results into three phase AC voltages generated on main machine stator. These AC 3-phase voltages are shown as A, B, C on the VFG. The VFG machine detailed diagram is shown as below.

Figure 5-57 Synchronous Machine Detailed Simulink Model

The two synchronous machine models have parameters for stator and rotor inductances and resistances, number of pole pairs etc. are selected as below.

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Figure 5-58 Exciter Machine and Main Machine Model Parameters

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Control Circuit The control circuit consists of the outer control loop which receives the 270VDc reference and 270VDC feedback and compares them to generate the error signal. The error signal passes through PI controller to generate the current reference signal. The inner control loop consists of the feed forward control which receives the load current reference signal (I_Load_Ref) and adds it to the actual load current (I_Load_Measured) measured at the output. The load current measured ahead of the filter capacitor (I_Load_Rectifer) is subtracted form this value to generate the error signal. So under steady state condition, both I_Load_Measured and I_Load_Rectifier are identical as capacitor does not draw any current. But during the transient conditions, I_Load_Measured responds faster than I_Load_Rectifier to improve the close loop control. Load Application Circuit:

150 Constant4

Repeating Sequence4 1 DC Load

Repeating Sequence3

Selector

Switch1

10 Constant3

Figure 5-58 Topology 3 Load Application Simulink Model

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Load application circuit shows that either 150 amps can e applied as a load all the time or by flipping the selector switch in lower position, load transients can be applied to study the response of the system.

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Output Voltage Build up- High Speed

Output Voltage Build Up

Figure 5-59 Topology 3 High Speed, Output Voltage build-up Output Voltage Ripple - High Speed

High Speed - Output Steady State Voltage Ripple – 15.6 kHz

Figure 5-60 Topology 3 High Speed Output voltage Ripple

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Load On and Load off Transients - High Speed

Load On Voltage Dip

Load Off Voltage Overshoot

Load Change 150A to 10A

Load Change 10A to 150A

Figure 5-61 Topology 3 High Speed Load-on & Load-off Transient

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VFG AC Voltage – 40kW DC on – High Speed

Figure 5-62 Topology 3 High Speed VFG AC Voltage Waveform

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VFG AC Current – 40kW DC on – High Speed

Figure 5-63 Topology 3 High Speed VFG AC Current Waveform 81

Exciter Field Stator Current – High Speed

Load On Load Off Initial Output Build Up

Figure 5-64 Topology 3 High Speed Exciter Field Current Waveform

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VFG Machine V and I Measurements – High Speed

Field Rotor Rectified V

Field Rotor I

Main Rotor I

Field Rotor L-L V

Field Stator I

Field Stator V

Figure 5-65 Topology 3 High Speed V, I Waveforms 83

Output Voltage Ripple - Low Speed

Low Speed - Output Steady State Voltage Ripple – 8.4 kHz

Figure 5-66 Topology 3 Low Speed Output voltage Ripple 84

Load On and Load Off Transients - Low Speed

Load On Voltage Dip

Load Off Voltage Overshoot

Load Change 10A to 150A

Load Change 150A to 10A

Figure 5-67 Topology 3 Low Speed Load-on & Load-off Transient 85

VFG 3-Phase AC Voltage Build Up, Load on Transient and Load off Transient

VFG Load On

VFG 3-Phase Steady State Voltage

VFG 3-Phase AC Voltage Build Up

VFG Load Off

Figure 5-68 Topology 3 High Speed Load-on & Load-off Transient VFG Side 86

Control loop Signal during Load Application–Low Speed

PID Loop#1 Error Signal

PID Loop#2 Current Reference

Load Current Feedback

Output Load Feed Forward

Bridge Current Feedback

PID Loop#2 Error Signal Feed Forward Effect Exciter Voltage Control Signal

Figure 5-69 Topology 3 Low Speed Control Loop Signals

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Output Voltage Dip- Load Application

Output Load Application 10A to 150A – Feed-Forward Control

Figure 5-70 Topology 3 Output Load on Feed-Forward

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Observations: Output Voltage AC Ripple The VFG with rectifier bridge topology provides 6-pulse regulated 270VDC output. Both main and Exciter machines have 6 pole pair, therefore the commutation frequency at low speed and high speed are estimated as below. f 14000 ( Hz ) =

Speed (rpm) * PP ( polepair ) 14000 * 6 = = 1400 Hz and 60 60

f 26000 ( Hz ) =

Speed (rpm) * PP ( polepair ) 26000 * 6 = = 2600 Hz 60 60

Therefore, low speed output voltage AC ripple has 8.4 kHz (1400 Hz * 6) frequency component in 1.2V peak-to-peak. High speed output voltage AC ripple has 15.6 kHz (2600 * 6) frequency component in 0.4V peak-to-peak. Output DC Voltage Transients At low speed, during the load application from 10A to 150A, the output DC voltage dips to ~225VDC and during the load removal from 150A to 10A, the output voltage overshoots to ~315VDC. At high speed, the output voltage dips to ~215VDC during the load application and overshoots to ~328VDC during the load removal. The higher voltage dip and overshoot at high speed compared to low speed is due to the higher voltage drop in the machine inductance at high frequency. Steady state voltage ripple and load transients are well within MIL-STD-704 specifications for power quality.

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Distortion Spectrum Low Speed: 300 0.7

FFT of Output VDC at 150 amp load

Magnitude (V)

200

0.5

150

270VDC

100

8400 Hz component = 0.715 Vpeak = 0.50Vrms

0.6

Magnitude (V)

250

0.4

0.3 0.2

50

0

0.1

0

0

0.2

0.4

0.6

0.8 1 1.2 Frequency (Hz)

1.4

1.6

1.8

2

7600

4

x 10

7800

8000 8200 Frequency (Hz)

Model Output VDC 271.2 271

8400

8600

8400 Hz component = 1.5Vpeak-topeak = 0.53Vrms

Output VDC

270.8 270.6 270.4 Output VDC 270.2 270 269.8 269.6 269.4 0.0526 0.0527 0.0528 0.0529 0.053 0.0531 0.0532 Time (ms)

Figure 5-71 Topology 3 Low Speed FFT of Output VDC The model output voltage ripple matches very close to FFT analysis.

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High Speed: 300 0.25

250

FFT of Output VDC at 150 amp load

Magnitude (V)

0.15

150

270VDC

100

0.1 0.05

14400 Hz component = 0.2 Vpeak = 0.14Vrms

0 -0.05

50

-0.1

0

0

0.2

0.4

0.6

0.8 1 1.2 Frequency (Hz)

1.4

1.6

1.8

2

1.4385

1.439

4

1.4395 Frequency (Hz)

x 10

Model Output VDC 271.1

8400 Hz component = 0.4Vpeak-topeak = 0.14Vrms

271 270.9 Output VDC

Magnitude (V)

200

0.2

270.8 270.7

Output VDC

270.6 270.5 270.4 270.3 0.055 0.0551 0.0552 0.0553 0.0554 0.0555 0.0556 Time (ms)

Figure 5-72 Topology 3 High Speed FFT of Output Voltage VDC

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1.44 4

x 10

The model output voltage ripple matches very close to FFT analysis.

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6.0 MEA (More Electric Aircraft) The ever increasing demand of reducing the operating and maintenance costs of the aircraft while having more and more power to be extracted from the engine demands a paradigm shift in a way how the traditional power available on the aircraft in terms of hydraulic, pneumatic and electric. The need to get more power extraction with lower cost is pushing the boundary to go towards more electric aircraft. The particular area of power generation and power conversion with the potential of achieving the excellence in fuel economy, higher power density is •

ESG (Engine Starter Generator) – ESG replaces conventional pneumatic starter and AC generator with integrated starter generator which can do both functions.

•

High Voltage DC Generating Systems – 270VDC high voltage DC systems are replacing conventional 115VAC systems to achieve significant feeder weight savings.

•

No Break power Transfer (NBPT) and Paralleling Operation with VFG systems.

•

Integrated Power generating System with high DC output.

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6.1 Power Converter Design Optimization High power density converter designs can be achieved by employing intelligent packaging concepts, modular power switches and state of art cooling designs. The PWB design using Thermal Clad material (Berquist Thermal) can replace the discrete and modular power switches. Thermal Clad is a cost effective solution which can eliminate components, allow more simplified designs, smaller devices and overall less complicated production processes. Additional benefits of Thermal Clad include lower operating temperatures, longer life and more durability. Power conversion applications can enhance their performance by replacing FR-4 with Thermal Clad dielectrics in multi-layer assemblies. Typical Thermal clad material has base layer of aluminum of copper, dielectric layer on top which has very low thermal resistance. The top of the dielectric layer is the circuit layer.

Circuit Layer

Dielectric Layer

Base Layer

This is the printed circuit foil with thickness of 1oz to 10oz (35-350µm) in standard Thermal Clad. This offers electrical isolation with minimum thermal resistance. The multiple-layer dielectric is the key element of Thermal Clad, and bonds the base metal and circuit metal together. The dielectric has UL recognition, simplifying agency acceptance of final assemblies This is often aluminum, but other metals such as copper may also be used. The most widely used base material thickness is 0.062" (1.6mm) in aluminum, although many thicknesses are available. In some applications, the base layer of metal may not be needed 94

Silicon carbide power devices have become more attractive alternate to silicon switches for certain applications demanding •

Higher operating temperatures – up to 300ºC

•

Lower On resistance for SiC based switches compared to Si switches.

•

Lower thermal resistance resulting into higher power density

•

Lower switching losses

•

Higher switching frequencies

SiC diodes are becoming more and more available but SiC MOSFETs are still few years away before it becomes economical resulting from process stabilization to achieve higher yield. In general, the switches for power converters can be broadly categorized as SCR, MOSFET and IGBTs. The switching frequency capability of the switches is listed below. •

SCR – 10 kHz

•

IGBT – 60 to 70 kHz

•

MOSFET - Up to 500kHz

The converter packaging design fundamentally is driven by the cooling mechanism either it is liquid cooled (such as PAO), self air cooled (fan), forced air cooled (external cooling duct or tubing) or convection/conduction cooled. The thermal modeling using the co-efficient of heat transfer and actual packaging of the components with their losses in watts at the worst case condition will help to identify if the design is capable

95

of handling the worst case environment (high temperature, sea level (ambient) or low temperature, high altitude (ambient) or high temperature, high altitude (controlled environment). This study becomes integral part of the risk mitigation to identify that all the switches selected can survive with sufficient thermal margin under worst case operating environment and worst case operating loads.

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7.0 Conclusions 1. The complete system level modeling including the power stage and control stage provides sufficient insights into the system level performance to ensure that the power quality requirements are met. This upfront modeling approach helps to create single iteration optimized designs. 2. The three topologies for the 270VDC converter system are modeled to compare the system performance and it is concluded that the two stage converter system with PFC front end followed by DC-DC converter provides the best power quality performance for the steady state regulation, ac ripple and load transients. This topology also provides the benefit of improving the power factor and reducing the distortion by using the d-q transformation for three phase voltages and currents. The d-q transformation helps to individually control the resistive and reactive power by employing independent control loops. 3. The system performance is better with using three phase fixed AC voltage source instead of the AC generator. This is because AC generator has internal voltage drop across the impedance which is resistance and reactance (n*f*LC), where,

97

caused by the winding

a. n=Number of pulses(n=6 for 6-pulse system) b. f=Frequency of the generator c. LC = Leakage inductance of the machine windings At high generator speed (high generator frequency), the internal impedance of the generator is high resulting into more voltage drop inside the machine. Therefore, at high speed, generator requires higher machine voltages for the same load compared to low speed. 4. The d-q control technique can not be employed in topology with the generator and 6-pulse rectifier-bridge. This is because the degree of freedom is limited to regulate the output voltages just by controlling the field current in absence of active switches such as MOSFETs or IGBTs. 5. The multi-pulse isolation transformer model in topology 2 with voltage and current sources approximates the actual transformer function by closely sharing the output DC current through each of the three phase secondary winding groups. 6. The converter topology 1 is a bidirectional converter so it can act as an inverter also. Therefore modeling it as a converter provides flexibility to make delta changes to simulate the model as an inverter. This topology can be really very helpful for engine starter generator applications where the bidirectional converter is in inverter mode to start the engine through machine. Once engine picks up speed it lights off and power flows from engine to machine to converter where converter converts 3-phase AC power to DC output to power the aircraft.

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7.1 Recommendation for Future Work Following work is considered in order to extend the full benefits of the thesis. •

Develop fully integrated power generation and conversion model with AC synchronous machine, AC-DC converter followed by DC-DC converter. This will pose a challenge to optimize the model due to added complexity of the machine model with the active power switches may slow down the simulation time.

•

Develop the bidirectional converter model using the AC-DC converter model and DC-AC inverter model to extend the thesis work for engine starter generator applications where the transition form the starter mode to generator mode will require bidirectional converter system.

• Extend the system simulation approach to develop the thermal model and identify the optimized weight vs. performance approach.

7.2 Thesis Contribution •

The thesis provides valuable insights into various topologies for DC converter systems for aerospace applications.

•

The modeling and simulation of the DC converter systems provide the baseline for the future designs by establishing the boiler plate.

•

The detailed theoretical explanation together with modeling for vector control and space vector modulation establishes strong foundation for understanding the modeling approach.

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•

The thesis provides the understanding of how to analyze the power quality of the output and identify the optimum approach for the design.

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[2]

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[3]

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[4]

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[5]

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[6]

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[7]

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[8]

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[10]

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[11]

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[12]

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[13]

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