## CONTROL OF DC MOTOR USING DIFFERENT CONTROL STRATEGIES

Transaction on Machine, Power electronics and Drives ISSN: 2229-8711 Online Publication, June 2011 www.pcoglobal.com/gjto.htm DC-P19/GJTO CONTROL OF ...
Author: Julian Malone
Transaction on Machine, Power electronics and Drives ISSN: 2229-8711 Online Publication, June 2011 www.pcoglobal.com/gjto.htm DC-P19/GJTO

CONTROL OF DC MOTOR USING DIFFERENT CONTROL STRATEGIES Hedaya Alasooly Elect. Eng., Bahrain University, Bahrain Email: [email protected] Received August 2010, Revised October 2010, Accepted November 2010

Control oOf DC Motor using Different Control Strategies

22

where, K f ω is a linear approximation for viscous friction. The electrical part of the motor equations can be described by K di R 1 (4) = − i − b ω + v app L L L dt Given the two differential equations, you can develop a statespace representation of the DC motor as a dynamic system. The current i and the angular rate are the two states of the system. The applied voltage, v app , is the input to the system, and the angular velocity

ω

U=Step

is the output. K ⎤ ⎡ R − − b⎥ i ⎡1⎤ d ⎡i⎤ ⎢ L L ⎡ ⎤ + ⎢ ⎥v = ⎢ ⎥ L app dt ⎢⎣ω ⎥⎦ ⎢ K m − K f ⎥ ⎢⎣ω ⎥⎦ ⎢ 0 ⎥ ⎣ ⎦ J ⎦⎥ ⎣⎢ J ⎡i⎤ y = [0 1]⎢ ⎥ + [0]v app ⎣ω ⎦

C

(5)

DC

Speed

(6) Fig. 2. Closed loop control of DC motor, C is the compensator

3. Controlling DC Motor Angular Velocity through Different Compensation Techniques In this paper the DC motor model was used in order to compare different control strategies and compensation techniques. The proposed control schemes were designed in order to derive the angular velocity ω to unity with best design criteria’s that can be achieved, i.e, rise time of less than 0.5 second, overshoot of less than 10%, gain margin greater than 20 dB, phase margin greater than 40 degrees The following nominal values for the various parameters of a DC motor used: R= 2.0 Ohm, L= 0.5 Henrys, Km = .015, Kb = .015, Kf = 0.2 , J= 0.02 kg·m², so the transfer function of the DC motor 1 1 ω ( s) (7) = = v app ( s ) s 2 + 14s + 40.02 ( s + 9.996)( s + 4.004)

ω ss = .035(rad / sec)

As the open loop step response has a large steady state error, ω ss = 0.037, ess = .963(rad / sec) , different closed loop control strategies and compensator designs were compared in this paper in order to eliminate the steady state error and enhance the system transient response. Fig. 2 shows the DC motor with negative unity feed back, and a feed forward compensator C added in series with the DC motor so it will control the applied voltage to DC motor. The main objective is to design a feed forward compensator C that will derive the DC motor angular velocity to unity.

t s ≈ 1.1(sec)

Fig1 shows the open loop response of the dc motor angular speed due to step input, vapp = 1, vap ( s) = 1 . s

3.1. Proportional Controller C=1 The motor was controlled with the feed forward proportional compensator C = 1 . The overall closed loop transfer function of the controlled system 1.5( s + 10)( s + 4) ω (s) (8) = U ( s ) ( s + 10)( s + 4)(s + 4.26) (s + 9.73) ωss = .036( rad / sec) t s ≈ 1.05(sec) Fig. 3 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system. It can be noted that the proportional compensator of C=1 could not reduce the steady state error in the angular speed. \3.2. Proportional Controller C=100: The motor was controlled with the feed forward proportional compensator C = 1 00. The overall closed loop transfer function of the controlled system ω ( s) 1.5( s + 10)( s + 4) (9) = U ( s) ( s + 10)(s + 4)(s + 4.26) (s + 9.73) ωss = .789(rad / sec) ts ≈ .58(sec)

0.04

Fig. 4 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system. It can be noted that the increasing the proportional compensator gain, C=100, reduced the steady state error in angular speed but could not eliminate it. The system will be stable as the propotional gain increased.

0.035 0.03

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Fig. 1. The open loop step response of the DC motor angular velocity ω (rad/sec) Copyright @ 2011/gjto

3.3. Integral Controller C = 100 / s : The motor was controlled with the feed forward integral compensator C = 100 / s . The overall closed loop transfer function of the controlled system 150s(s + 10)(s + 4) ω(s) (10) = U(s) s(s + 10)(s + 4) (s + 11)(s+ 1.16+ 3.4i)(s+ 1.16- 3.4i) ωss = 1(rad / sec) t ss ≈ 5(sec)

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Fig. 3. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C=1. Fig. 5 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system. It can be noted that the integral controller eliminated the steady state error in angular speed so the steady state response is improved, but the settling time and amount of the overshoot are large, also the system is subject to instability problems as the integral gain increased, so a compensator consisting of an integrator is not enough to satisfy the design requirements. 3.4. Proportional Integral Controller C = 100 * (1 + s)/s : The motor was controlled with the feed forward proportional integral compensator C = 100 * (1 + s)/s . The overall closed loop transfer function of the controlled system ω(s) 150s(s +10)(s + 4)(s +1) (11) = U(s) s(s +10)(s + 4) (s + 0.8)(s+ 6.58+11i)(s+ 6.58−11i) ωss = 1(rad / sec) tss ≈ 5(sec) Fig. 6 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system. It can be noted hat the proportional and integral controller eliminated the steady state error, the system is stable as the controller gain increased, the amount of overshoot is reduced, but the system settling time still high. Copyright @ 2011/gjto

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Fig. 4. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C=100. 3.5. Phase Lag Compensator C = 100 * (s + 0.1)/(s + 0.01) : Generally the lag compensator has the following form, A s + 1 / T , is known to result in large increase in gain C= α s + 1 /(αT ) which means a much smaller steady state error, and a decrease in ωn and so has the disadvantage of producing an increase in settling time. The zero s = −1 / T and the pole s = −1 /(αT ) are selected close together with α is chosen large value such as 10. The pole and zero are located to the left and close to origin, these results in increased gain. The motor was controlled with the feed forward phase compensator, C = 100 * (s + 0.1)/(s + 0.01) . The overall closed loop transfer function of the controlled system 150(s + 10)(s + 4)(s + 0.1)(s + 0.01) ω(s) = U(s) (s + 10)(s + 4) (s + 0.08)(s+ 0.01)(s+ 6.9 +11.8i)(s+ 6.9 - 11.8i)

ωss = .974(rad / sec)

ts ≈ 27.4(sec)

(12)

Fig. 7 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system. It can be noted that the steady state error is reduced but not fully eliminated while the settling time is large. The controlled

Control oOf DC Motor using Different Control Strategies

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Fig. 5. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C = 100 / s . system is not subject to instability problem as the controller gain increased. 3.6. Derivative Compensator C = s : The motor was controlled with the feed forward derivative compensator C = s . The overall closed loop transfer function of the controlled system 0.5s ( s + 10)( s + 4) ω (s) (13) = U ( s ) ( s + 10)( s + 4) (s + 12.23)(s + 3.273) ωss = 0 t s ≈ 1.44

Fig. 8 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system..It can be noted that derivative compensator will derive the motor angular speed to zero and so the steady state error is not acceptable. 3.7. Lead Integral Compensator C = 100 * (s + 10)/(s(s + 100)) settling time. The zero s = −1 / T is superimposed on a pole of the original system, and that results in moving the root locus to left and thus increasing the undamped natural frequency. α = 0.1 is a common choice. Generally the lead compensator has the following form: Copyright @ 2011/gjto

P.M.: 61.4 deg Freq: 9.92 rad/sec

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Fig. 6. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C = 100 * (1 + s)/s . s + 1 / T . The lead compensator results in moderate s + 1 / αT increase in gain and thereby improving the steady state error. It also results in large increase in ωn and therefore reduces the C=A

The motor was controlled with the feed forward lead and integral compensator C = 100 * (s + 10)/(s(s + 100)) . The overall closed loop transfer function of the controlled system ω(s) 150s(s + 10)2 (s + 4)(s + 100) U(s)

=

s(s + 10)2 (s + 4) (s + 3.56)(s + 3.27)(s + 100)2 (s + 0.42)

ωss = 1(rad / sec)

t ss ≈ 15(sec)

(14)

Fig. 9 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system. It can be noted that the lead integral compensator will eliminate the steady state error, but the transient response settling time is large, also the system is subject to instability problems as the controller gain increased. 3.8. Lead Lag Compensator C = 100( s + 10)( s + 0.1) /(( s + 100)( s + 0.01)) Lead lag compensator shall combine the desirable characteristic of the lead and lag compensators. It shall result in large increase

Control oOf DC Motor using Different Control Strategies

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in gain which improves the steady state response, and it shall result in an increase in ωn , which improves the transient settling time. The motor was controlled with the feed forward lead and integral compensator C = 100( s + 10)(s + 0.1) /((s + 100)(s + 0.01)) . The overall closed loop transfer function of the controlled system 150(s +100) (s +10)(s +10)(s + 4)(s + 0.1)(s + 0.01) (s +100) (s + 98.4)(s +10)(s +10) (s + 5.57)(s+ 4)(s+ 0.035) (s + 0.01)

t ss ≈ 101(sec)

(15)

Fig. 10 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system. It can be noted that the lead lag compensator will only reduce the steady state error, while the transient response settling time is very large. The controlled system is not subject to instability problem as the controller gain increased. 3.9. Proportional Integral Derivative Compensator (PID) C = 100 + 100 / s + 100 * s The motor was controlled with the feed forward PID compensator C = 100 + 100 / s + 100 * s . The overall closed loop transfer function of the controlled system

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Fig. 7. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C = 100 * (s + 0.1)/(s + 0.01) .

ωss = 0.78(rad / sec)

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Fig. 8. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C = s

ω (s)

150s ( s + 10)( s + 4)( s 2 + s + 1) U ( s ) s( s + 162.8)( s + 10)( s + 4) (s 2 + 1.16s + 0.92) (16) ωss = 1(rad / sec) t ss ≈ 8(sec) =

Fig. 11 shows the closed loop step response of the DC motor angular velocity ω , the root locus and bode plots of the controlled system. It can be noted that the PID compensator can eliminate the steady state error, but still the transient response settling time is quite large. The controlled system will have poles in the imaginary axis as the controlled gain increased. 4. Linear quadratic tracker design: The continuous linear quadratic tracker problem [2] is summarized as follows. The system model, x& = f ( x, y ) = Ax + Bu + Ed (17)

y = Cx + Du + Fd

To

keep

a

specified

(18) linear combination of the states

y = Cx + Du + Fd close to given reference track r (t ) , let us

prescribe the quadratic cost index,

Control oOf DC Motor using Different Control Strategies

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Fig. 9. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C = 100 * (s + 10)/(s(s + 100))

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Fig. 11. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C = 100 + 100 / s + 100 * s

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J = ϑ( x(T ), T ) + ∫ [L( x, u, t )dt

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T 1 [(Cx + Du + Fd − r) Q(Cx + Du + Fd − r )] +[u T Ru]dt 2 to∫

If we define the Hamiltonion function,

H = L( x, u, t ) + λT f ( x, u, t )

0

(20)

L = [Cx + Du − r]T Q[Cx + Du − r] + uT Ru

-0.5 -45

L = (Cx)T Q(Cx) + (Cx)T Q(Du) + (Cx)T Q(Fd) − (Cx)T Qr + (Du)T Q(Cx) + (Du)T Q(Du) + (Du)T Q(Fd )

-90

−(Du)T Qr − rT Q(Cx) − rT Q(Fd ) − rT Q(Du) + rT Qr + uT Ru + (Fd )T Q(Cx) + (Fd )T Q(Du)

-1

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(19)

1 J = [(Cx(T ) + Du(T ) + Fd(T ) − r (T ))T 2 F (Cx(T ) + Du(T + Fd(T ) − r (T ))]

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−(Fd)T Qr + (Fd)T Q(Fd)

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Fig. 10. The closed loop step response of the DC motor angular velocity ω , the root locus and bode plot of the controlled system when C = 100( s + 10)( s + 0.1) /(( s + 100)( s + 0.01)) Copyright @ 2011/gjto

The optimal control is given by solving, State system,

∂H = f (x, u, t) = Ax + Bu + Ed ∂λ y = Cx + Du + Fd

x& =

t ≥ t0 (21) (22)

Control oOf DC Motor using Different Control Strategies Costate system, ∂H ∂f ∂L −λ& = = ( )T λ + t ≤T ∂x ∂x ∂x −λ& = ATλ + (CTQC)x + (CTQD)u + (CTQF)d −CTQr

(23) t ≤T

Stationary conditions,

0=(

∂f T ∂L ) λ +( ) ∂u ∂u

(24)

∂L = Ru + DT QCx − DT Qr + DT QDu + DT QFd (25) ∂u 0 = BTλ + Ru + DTQCx − DTQr + DTQDu + DTQFd (26) u = −(R + DTQD)−1(DTQCx − DTQr + DTQFd + BT λ) (27) Then, the optimal controller becomes,

x& = Ax + Bu + Ed

t ≥ t0

(28)

− λ& = AT λ + (CT QC) x + (CT QD)u − CT Qr

(29)

−1

T

T

T

v& = [H4 − SH2 ]v + [H6 − SH5 ]r + [H8 − SH7 ]d (43) S& = H − SH + H S − SH S (44) 3

1

4

2

− S& = − H 3 + SH 1 − H 4 S + SH 2 S

(45)

AoT K + KAo + Qo − KBRo−1 B T K = 0

(46)

Where,

H 1 = Ao = A − BRo−1 D T Q C

(47)

−1 o

A = − H 4 = A − C QDR B T o

T

T

T

(48)

Q0 = − H 3 = C T QC − C T QDRo−1 D T QC

(49)

H 2 = − BRo−1 B T

(50)

.

In steady state, v = 0 v = −[H4 − SH2 ]−1[H6 − SH5 ]r +[H4 − SH2 ]−1[H8 − SH7 ]d (51)

v = Kr r + Kd d

Thus, −1

u = −(R + D QD) (D QCx − D Qr + B λ + D QFd ) (30) T

27

T

u = − Ro [ BT (Sx + v) + DT QCx − DT Qr + DT QFd ] u = −Ro−1 (B T S + DT QC) x +

−λ& = AT λ + (C T QC ) x + (C T QD)u − C T Qr + C T QFd

(−Ro−1 B T K r + Ro−1 DT Q)r − ( R0−1 B T K d + Ro−1 DT QF)d

= ( AT − C T QDRo−1 BT )λ + (C T QC − C T QDRo−1 DT QC ) x

(52)

+(C T QDRo−1 DT Q − C T Q)r + (−C T QDRo−1 DT QF + C T QF )d

We can summarise that continuous linear quadratic tracker optimal control as follows, u = Fx x + Fr r + Fd d (53)

( 31) If we considered,

−1

Fx = − Ro ( BT S + DT QC )

H 1 = A − BRo−1 D T QC

(32)

H 2 = − BRo−1 B T

(33)

H 3 = −(C T QC − C T QDRo−1 D T QC )

(34)

Fd = − Ro−1 BT K d − Ro−1 DT QF

H 4 = −( AT − C T QDRo−1 B T )

(35)

Where, S is the solution of the Riccati equation

H 5 = BR D Q

(36)

AoT S + SAo + Qo − SBRo−1 B T S = 0

(55)

H 6 = −(C T QDRo−1 D T Q − C T Q)

(37)

H 1 = Ao = A − BRo−1 D T Q C

(56)

H 7 = E − BRo−1 DT QF

(38)

AoT = − H 4 = AT − C T QDRo−1 B T

(39)

Q0 = − H 3 = C QC − C QDR D QC

(58)

Ro = R + D QD

(59)

H 1 = A − BRo−1 D T QC

(60)

−1 o

T

H 8 = −C T QDRo−1 DT QF + C T QF

Fr = − Ro−1 BT K r + Ro−1 DT Q

T

(54)

−1 o

T

(57) T

T

Then

⎡ x& ⎤ ⎡ H1 ⎢ λ& ⎥ = ⎢ H ⎣ ⎦ ⎣ 3

H 2 ⎤ ⎡ x ⎤ ⎡ H5 ⎤ ⎡H ⎤ + ⎢ ⎥ r + ⎢ 7 ⎥ d (40) ⎥ ⎢ ⎥ H 4 ⎦ ⎣λ ⎦ ⎣ H 6 ⎦ ⎣ H8 ⎦

From that, we have Copyright @ 2011/gjto

(61) −1 o

H 3 = −(C QC − C QDR D QC ) T

T

−1 o

T

(62)

H 4 = −( A − C QDR B )

(63)

(41)

H 5 = BRo−1 D T Q

(64)

(42)

H 6 = −(C T QDRo−1 D T Q − C T Q)

Substituting,

λ = Sx + v λ& = S&x + Sx& + v&

H 2 = − BRo−1 B T T

T

−1 o

H 7 = E − BR D QF T

T

(65) (66)

Control oOf DC Motor using Different Control Strategies

28 H 8 = −C T QDRo−1 DT QF + C T QF

References

(67)

After the solution of the linear quadratic tracker problem, the following control scheme is applied

Vapp = k1i + k 2ω + k 3 * 1

Fig. 12 shows the closed loop step response of the DC motor angular velocity ω . ωss = 1(rad / sec) t ss ≈ 1(sec) . So, the designed linear quadratic has the best steady state and transient responses. It fully eliminated the steady state error with small transient settling time. There is no overshoot and the system is completely stable. 1 0.9 0.8 0.7

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2

Fig. 12. The closed loop step response of the DC motor angular velocity ω when the optimal output tracker applied 5. Conclusion: A simple model of a DC motor driving an inertial load has the angular rate of the load, ω , as the output and applied voltage,

v app , as the input. Different control strategies and compensator designs with the objective to control the angular speed to be unity with the best steady state and transient performance. The comparision was made between the proptional controller, integral controller, propotional and integral controller, phase lag compensator, derivitive controller, lead integral compensator, lead lag compensator, PID controller and the linear quadratic tracker design based on the optimal control theory. It was found that the designed linear quadratic gave the best steady state and transient responses performances. It fully eliminated the steady state error with the least transient settling time. There is no overshoot and the system is completely stable. The reason is that the other compensator designs are mostely based on trial and error while the linear quadratic tracker design is based on the optimal control theory which can give best dynamic performance for the controlled system.